Power converters with modular stages

ABSTRACT

An apparatus for controlling a power converter that includes an inductance and a switched-capacitor network that cooperate to transform a first voltage into a second voltage features a controller, a switched-capacitor terminal for connection to the switched-capacitor network, and switches. at least one of which connects to the switched-capacitor terminal.

RELATED APPLICATIONS

This application is a continuation of U.S. application Ser. No.16/085,680, filed on Sep. 17, 2018, now U.S. Pat. No. 10,381,924, whichis a national phase under 35 USC 371 of international application no.PCT/US2017/023191 filed Mar. 20, 2017, which claims the benefit of theMar. 18, 2016 priority date of U.S. Provisional Application 62/310,235.In addition, international application no. PCT/US2017/023191 is acontinuation-in-part under 35 USC 120 of Ser. No. 15/138,692, filed onApr. 26, 2016, now U.S. Pat. No. 9,712,051, which is a continuation ofU.S. application Ser. No. 14/513,747, filed on Oct. 14, 2014, now U.S.Pat. No. 9,362,826, which is a continuation of U.S. application Ser. No.13/771,904, filed on Feb. 20, 2013, now U.S. Pat. No. 8,860,396, whichis a continuation of International Application No. PCT/US2012/036455,filed on May 4, 2012, which claims the benefit of the priority date ofU.S. Provisional Application Nos. 61/482,838, filed on May 5, 2011; U.S.Provisional Application No. 61/548,360, filed on Oct. 18, 2011; and U.S.Provisional Application No. 61/577,271, filed on Dec. 19, 2011. Thecontent of these applications is hereby incorporated by reference in itsentirety.

FIELD OF DISCLOSURE

This disclosure relates to power supplies, and in particular to powerconverters.

BACKGROUND

Many power converters include switches and one or more capacitors thatare used, for example, to power portable electronic devices and consumerelectronics. Switch-mode power converters regulate the output voltage orcurrent by switching energy storage elements (i.e. inductors andcapacitors) into different electrical configurations using a switchnetwork.

Switched-capacitor converters are switch-mode power converters thatprimarily use capacitors to transfer energy. These converters transferenergy from an input to an output by using switches to cycle a networkof capacitors through different topological states. A common converterof this type, known as a “charge pump,” is commonly used to produce thehigh voltages in FLASH memories and other reprogrammable memories.Charge pumps have also been used in connection with overcoming thenuclear strong force to transform one element into another.

In a switched-capacitor converter, the number of capacitors and switchesincreases as the transformation ratio increases. Switches in the switchnetwork are usually active devices that are implemented withtransistors. The switch network may be integrated on a single or onmultiple monolithic semiconductor substrates, or formed using discretedevices. Furthermore, since each switch in a power converter normallycarries high current, it may be composed of numerous smaller switchesconnected in parallel.

SUMMARY

Typical DC-DC converters perform voltage transformation and outputregulation. This is usually done in a single-stage converter such as abuck converter. However, it is possible to split these two functionsinto two specialized stages, namely a transformation stage, such as aswitching network, and a separate regulation stage, such as a regulatingcircuit. The transformation stage transforms one voltage into another,while the regulation stage ensures that the voltage and/or currentoutput of the transformation stage maintains desired characteristics.

In one aspect, the invention features an apparatus for controlling apower converter having an inductance and a switched-capacitor networkthat are connected to transform a first voltage into a second voltage.Such an apparatus includes a switched-capacitor terminal for connectionto the switched-capacitor network, and switches, at least one of theswitches being connected to the switched-capacitor terminal. Theapparatus also includes a controller that is connected to both theregulating circuit and to the switches. The controller's structure issuch that it causes the inductance and the switched-capacitor network tocooperate in causing transformation of the first voltage into the secondvoltage. In some but not all embodiments, the inductance is aconstituent of a regulating circuit.

In some embodiments, the power converter includes a diode circuitcomprising first, second, third, and fourth diodes, with a cathode ofthe third diode and an anode of the second diode meeting at a firstnode, cathodes of the first and second diodes meeting at a second node,an anode of the first diode and a cathode of the fourth diode meeting ata third node, and anodes of the third and fourth diodes meeting at afourth node. In such embodiments, the first and third nodes areconfigured to be connected to an AC source and the second and fourthnodes are connected to the power converter.

In some embodiments, the controller comprises a first control circuitand a second control circuit, the first and second control circuitsbeing isolated from each other. In some embodiments, the controlcircuits are in galvanic isolation relative to each other. Also amongthe embodiments are those in which the first and second control circuitsare magnetically isolated from each other, those in which they areelectrically isolated from each other, and those in which they areinductively isolated from each other.

Embodiments also include those in which the controller has a firstcontrol circuit and a second control circuit, the first and secondcontrol circuits being isolated from each other but with additionalstructure that enables them to communicate optically, throughelectromagnetic waves, mechanically, through sound waves, and throughstatic and quasi-static electric and/or magnetic fields.

Other embodiments include at least one integrated circuit. Theseembodiments include those in which the control has a first controlcircuit and a second control circuit that are part of the sameintegrated circuit. In those embodiments that have two or moreintegrated circuits, there are embodiments in which first and secondcontrol circuits of the controller are in different ones of theintegrated circuits.

Also among the embodiments in which the controller has first and secondcontrol circuits are those in which the two control circuits outputcorresponding first and second control signals with the first controlsignal being a voltage difference between a first voltage and a secondvoltage that is lower than the first voltage, and the second controlsignal being a voltage difference between a third voltage and a fourthvoltage that is both lower than the third voltage and different from thesecond voltage.

Also among those embodiments in which the controller has first andsecond circuits are those in which the first and second control circuitsoutput corresponding first and second control signals that lack a commonground.

In yet other embodiments, the power converter further comprises aninductance connected to the switched-capacitor network for constraininginter-capacitor charge transfer within the switched-capacitor network.

Also among the embodiments are those in which the power converterfurther comprises a non-capacitive element connected to theswitched-capacitor network for constraining inter-capacitor chargetransfer within the switched-capacitor network.

In some embodiments, the controller is configured operate the switchingnetwork to cause the switched-capacitor network to transition betweenany two of at least three switching arrangements.

In other embodiments, the controller is configured to reconfigure theswitched capacitor network during operation thereof.

In some embodiments, the power converter includes a bridge rectifierconfigured to be connected to an AC source. In still other embodiments,the controller is configured operate the switching network to cause theswitched-capacitor network to transition between three states, whereinin a first state, power is supplied by a first set of capacitors in theswitched-capacitor network, wherein in a second state, power is suppliedby a second set of capacitors in the switched-capacitor network, and ina third state between the first and second states, no power is beingsupplied from the switched-capacitor network.

Also among the embodiments are those in which the controller isconfigured operate the switching network in multi-phase mode.

Some embodiments further include a diode circuit comprising first,second, third, and fourth diodes, wherein a cathode of the third diodeand an anode of the second diode meet at a first node, wherein cathodesof the first and second diodes meet at a second node, wherein an anodeof the first diode and a cathode of the fourth diode meet at a thirdnode, wherein anodes of the third and fourth diodes meet at a fourthnode, wherein the first and third nodes are connected to an AC source,and wherein the second and fourth nodes are connected to the powerconverter.

Additional embodiments include those in which a circuit that receives aninput AC voltage and an input AC current separated by a first phaseangle, and that outputs an output AC voltage and an output AC currenthaving a voltage and a current that are in phase.

In other embodiments, the controller comprises first and second controlcircuits that connect to different sides of a transformer.

Other embodiments feature a diode circuit and a filter circuit. In theseembodiments, the diode circuit comprises first, second, third, andfourth diodes, wherein a cathode of the third diode and an anode of thesecond diode meet at a first node, wherein cathodes of the first andsecond diodes meet at a second node, wherein an anode of the first diodeand a cathode of the fourth diode meet at a third node, wherein anodesof the third and fourth diodes meet at a fourth node, wherein the firstand third nodes are connected to an AC source. The filter circuit, onthe other hand, is configured to filter high-order harmonics of the ACsource, thereby suppressing radiation.

Yet other embodiments feature an AC bridge circuit connected between anAC source and the power converter.

Still other embodiments include a power-factor correction circuitconnected to the power converter.

Also among the embodiments are those that include an EMI filter at thepower converter.

These and other features of the invention will be apparent from thefollowing detailed description and the accompanying figures, in which:

DESCRIPTION OF THE FIGURES

FIG. 1 shows a power converter with a separable transformation stage andregulation stage;

FIG. 2 shows a power converter similar to that shown in FIG. 1 but withan isolated transformation stage;

FIGS. 3 to 10 show different ways of connecting transformation andregulation stages;

FIG. 11 shows a DC-DC converter with a separate regulating circuit andswitching network;

FIG. 12 explicitly shows control circuitry associated with a converteras shown in FIG. 11;

FIG. 13 shows details of the control circuitry shown in FIG. 12;

FIG. 14 shows signals present during operation of the control circuitryof FIG. 13.

FIG. 15 is a close-up of four signals from FIG. 14 showing the dead-timeinterval;

FIG. 16 shows details of switch layout in a converter similar to thatshown in FIG. 1;

FIGS. 17 and 18 show dependence of switching period and peak-to-peakripple as a function of output load current in two embodiments of thecontrol circuitry as shown in FIG. 12;

FIG. 19 shows a multi-phase converter similar to that shown in FIG. 12;

FIGS. 20 and 21 show signals present during operation of the controlcircuitry of FIG. 19;

FIG. 22 shows a bidirectional version of FIG. 11;

FIGS. 23-24 show DC-DC converters with alternate configurations ofregulating circuits and switching networks;

FIG. 25 shows a DC-DC converter like that shown in FIG. 24 with acontroller;

FIG. 26 shows another configuration of a DC-DC converter;

FIG. 27 shows a particular implementation of the power converterillustrated in FIG. 26;

FIG. 28 shows an embodiment with multiple regulating circuits;

FIG. 29 shows an RC circuit;

FIG. 30 shows a model of a switched capacitor DC-DC converter;

FIG. 31 shows an isolated variant of FIG. 30;

FIG. 32 shows output resistance of a switched-capacitor network as afunction of switching frequency;

FIGS. 33-34 show a series-parallel SC converter operating in chargephase and discharge phase respectively;

FIG. 35 shows a series pumped symmetric cascade multiplier with diodes;

FIG. 36 shows a parallel pumped symmetric cascade multiplier withdiodes;

FIG. 37 shows charge pump signals;

FIG. 38 shows a two-phase symmetric series pumped cascade multiplierwith switches;

FIG. 39 shows a two-phase symmetric parallel pumped cascade multiplierwith switches;

FIG. 40 shows four different cascade multipliers along withcorresponding half-wave versions;

FIG. 41 shows the circuit of FIG. 29 with an auxiliary converter used toreduce loss associated with charging a capacitor;

FIG. 42 shows an implementation of the circuit of FIG. 41;

FIG. 43 shows a cascade multiplier with clocked current sources;

FIG. 44 shows output impedance of a switched-capacitor converter as afunction of frequency;

FIGS. 45, 46, and 47 show clocked current sources;

FIG. 48 shows a cascade multiplier with the clocked current source ofFIG. 46;

FIG. 49 shows a particular implementation of the DC-DC converterillustrated in FIG. 22 with a full-wave adiabatically charged switchingnetwork;

FIG. 50 shows the DC-DC converter illustrated in FIG. 48 during phase A;

FIG. 51 shows the DC-DC converter illustrated in FIG. 48 during phase B;

FIG. 52 shows various waveforms associated with a 4:1 adiabaticallycharged converter;

FIG. 53 shows adiabatic charging of series connected stages;

FIG. 54 shows a particular implementation of the power converterillustrated in FIG. 53;

FIG. 55 shows an AC-DC power converter architecture;

FIG. 56 shows an AC voltage rectified using a reconfiguredswitched-capacitor stage;

FIG. 57 shows an embodiment of the AC-DC power converter architecture inFIG. 55, which includes an AC switching network;

FIG. 58 shows a particular implementation of the AC-DC converterillustrated in FIG. 57;

FIG. 59 shows the AC-DC converter illustrated in FIG. 58 during thepositive portion of the AC cycle;

FIG. 60 shows the AC-DC converter illustrated in FIG. 58 during thenegative portion of the AC cycle;

FIG. 61 shows an AC-DC power converter architecture with power-factorcorrection;

FIG. 62 shows a converter having an isolated controller;

FIG. 63 shows an alternative architecture of the converter in FIG. 62where the switching network is loaded by an LC filter;

FIG. 64 shows a converter in which a control signal for the regulatingcircuit is isolated from a control signal for the switching network;

FIG. 65 shows a configuration of FIG. 23 with an isolated controller asshown in FIG. 64;

FIG. 66 shows a configuration of FIG. 26 with an isolated controller asshown in FIG. 64;

FIG. 67 shows an implementation of the rectifier shown in FIG. 55;

FIG. 68 shows an alternative implementation of the rectifier shown inFIG. 55;

FIG. 69 shows an implementation of an EMI filter from the rectifiersshown in FIGS. 67 and 68;

FIG. 70 shows an alternative implementation of an EMI filter from therectifiers shown in FIGS. 67 and 68;

FIG. 71 shows an implementation of an AC bridge for use in theembodiments shown in FIGS. 67 and 68;

FIG. 72 shows one transformation stage driving two parallel regulationstages;

FIGS. 73 and 74 show particular implementations of the DC-DC converterillustrated in FIG. 22;

FIGS. 75 and 76 show particular implementations of the DC-DC converterillustrated in FIG. 24;

FIGS. 77 and 78 show particular implementations of the DC-DC converterillustrated in FIG. 23;

FIGS. 79 and 80 show particular implementations of the DC-DC converterillustrated in FIG. 26;

FIG. 81 shows a switching network implemented as a stack of layers;

FIGS. 82-85 are cross-sections of the stack in FIG. 81 with differentorders of passive and active layers;

FIGS. 86-89 show different locations of active and passive device facesfor the two-layer stack shown in FIG. 82;

FIGS. 90-93 show different locations of active and passive device facesfor the two-layer stack shown in FIG. 83;

FIG. 94 shows an implementation of FIG. 82 in which the passive devicelayer has a planar capacitor;

FIG. 95 shows an implementation of FIG. 82 in which the passive devicelayer has a trench capacitor;

FIG. 96 shows an implementation of FIG. 94 with wafer-to-wafer bondinginstead of die-to-die bonding;

FIG. 97 shows an implementation of FIG. 96 but with the device face ofthe active layer being its upper face instead of its lower face;

FIG. 98 shows three partitioned current paths of a switching network;

FIG. 99 shows an active layer with eight switches superimposed on eightcapacitors on a passive layer below it;

FIG. 100 shows one of the switches in FIG. 99 that has been partitionedinto nine partitions;

FIG. 101 shows a divided switching but not partitioned switch andcapacitor;

FIG. 102 shows a partitioned switch and capacitor;

FIG. 103 shows a capacitor partitioned in two dimensions; and

FIG. 104 shows a travel adapter having a power converter.

DETAILED DESCRIPTION

Some power converters carry out both regulation and transformation witha limited number of circuit components by comingling these functionsinto a single stage. As a result, certain components are used both forregulation and transformation. Sometimes the regulation stage isreferred to as a regulating circuit and the transformation stage isreferred to as a switching network. As used herein, they are equivalent.

FIG. 1 shows a modular multi-stage power converter that separates theconverter's transformation and regulation functions. These functions areno longer accomplished together as they would be in a single-stageconverter design. As a result, in a multi-stage power converter, asshown in FIG. 1, it is possible to optimize a transformation stage and aregulation stage for their specific functions. The transformation stageand the regulation stage can be treated as either independent entitiesor coupled entities.

In the power converter of FIG. 1, a transformation stage receives aninput voltage V_(IN) across its two input terminals and outputs anintermediate voltage V_(X) across its two output terminals at a fixedvoltage conversion ratio. Therefore, the intermediate voltage V_(X)changes in response to changes in the input voltage V_(IN). Thetransformation stage is thus regarded as “variable” if the voltageconversion ratio can be varied. However, it is not required that atransformation stage be “variable”.

In the particular embodiment shown in FIG. 1, there exists an electricalconnection between the transformation stage's negative input terminaland its negative output terminal. In this configuration, thetransformation stage is said to be “non-isolated.” In contrast, theembodiment shown in FIG. 2, no such connection exists between thetransformation stage's negative input and its negative output. Anexample of such a transformation stage is shown in FIG. 31 with avoltage conversion ratio of N₁:N₂.

In general, two functional components of a circuit or system are said tobe isolated, in a galvanic sense, if no direct conduction path existsbetween those two components, and yet energy and information can stillbe communicated between those components. The communication of suchenergy and information can be carried out in a variety of ways that donot require actual current flow. Examples include communication viawaves, whether electromagnetic, mechanical, or sonic. Electromagneticwaves in this context include waves in the visible range, as well asjust outside the visible range. Such communication can also beimplemented via static or quasi-static electric or magnetic fields,capacitively, inductively, or by mechanical means.

Galvanic isolation is particularly useful for cases in which the twofunctional components have grounds that are at different potentials.Through galvanic isolation of components, it is possible to essentiallyforeclose the occurrence of ground loops. It is also possible to reducethe likelihood that current will reach ground through an unintendedpath, such as through a person's body.

The transformation stage efficiently provides an intermediate voltageV_(X) that differs from the input voltage V_(IN) and that varies over amuch smaller range than the input voltage V_(IN). In practice, theintermediate voltage V_(X) varies during operation if there are changesat either the input or output of the transformation stage. Thesevariations require correction to achieve the desired output voltageV_(O). It is for this reason that a regulation stage is necessary. Asshown in FIGS. 1 and 22, a regulation stage receives the intermediatevoltage V_(X) across its input terminals and provides a regulatedvoltage V_(O) across its output terminals.

The architecture shown in FIG. 1 is flexible enough to permit designswith different requirements. For example, if magnetic isolation isrequired, a magnetic isolated fly-back converter can be used. Designsthat require multiple regulated output voltages can be accomplished byusing two separate regulation stages and a single transformation stage.

The architecture shown in FIG. 1 in effect creates a modulararchitecture for power converters in which fundamental building blockscan be mixed and matched in a variety of ways to achieve particulargoals.

FIGS. 3-10 are block diagrams showing different ways to arrange thetransformation stage and the regulation stage relative to a source or aload. The fact that these can even be represented as block diagrams atall stems from the modularity of the architecture. Such modularity isnot present in a conventional single-stage converter. In such aconverter, the functions of regulation and transformation are sointimately comingled that it is not possible to extract two separatecircuits and to say that one carries out regulation and the othercarries out transformation. Instead, in a conventional converter, if oneattempts to extract two circuits, one of which is a regulator and theother of which is a voltage transformer, the usual result is twocircuits that do not work.

FIG. 3 shows a generic architecture in which a pair of transformationstages sandwiches a regulation stage. Each transformation stage includesone or more switched-capacitor networks. Similarly, each regulationstage includes one or more regulating circuits. It is also possible tohave more than one source and more than one load. The double-headedarrows in FIG. 3 and in other figures indicate bidirectional power flow.

FIG. 4 shows a source-regulating configuration in which power flows froma source to a transformation stage. The transformation stage thenprovides the power to a regulation stage, which then passes it to aload. Thus, in this configuration, the load ultimately receives powerfrom the regulation stage.

In contrast, FIG. 5 shows a load-regulating configuration. In aload-regulating configuration, power flows from a source to a regulationstage, which then regulates it and passes it to a transformation stage.In this embodiment, the load receives power directly from thetransformation stage instead of directly from the regulation stage.

FIG. 6 shows a reverse source-regulating configuration similar to thatshown in FIG. 4, but with power flowing in the opposite direction.

FIG. 7 shows a reverse load-regulating configuration similar to thatshown FIG. 5, but with power flowing in the other direction.

In the embodiments shown in FIGS. 8 and 9, two transformation stagesbracket a regulation stage. These are distinguished by direction ofcurrent flow. FIG. 8 shows a source/load-regulating configuration inwhich power flows from the source to the load via a first transformationstage, a regulation stage, and a second transformation stage, and FIG. 9shows a reverse source/load-regulating configuration in which powerflows from the load to the source via a first transformation stage, aregulation stage, and a second transformation stage.

In another embodiment, shown in FIG. 10, several regulating circuitsrely on the same switched-capacitor converter. Note that of the threepower paths, a first and second power path are in the load-regulatingconfiguration whereas the third power path is in thesource/load-regulating configuration. An embodiment having severalregulating circuits is particularly useful since it enables differentoutput voltages to be provided to different loads.

FIG. 11 shows a modular DC-DC converter 10 assembled by combining twomodules using the principles suggested by FIG. 1. The illustratedmodular DC-DC converter 10 includes a switching network 12A thatconnects to a voltage source 14 at an input end thereof. An input of aregulating circuit 16A connects to an output of the switching network12A. A load 18A connects to an output of the regulating circuit 16A.Power flows between the voltage source 14 and the load 18A in thedirection indicated by the arrows. To simplify representation, theseparation of the connection into positive and negative lines has beenomitted.

The various configurations shown above have switches that need to beopened and closed at certain times. Thus, they all implicitly require acontroller to provide control signals that open and close theseswitches. The structure and operation of such a controller 20A isdescribed in connection with FIGS. 12-21.

FIG. 12 shows the modular DC-DC converter 10 of FIG. 11, but with acontroller 20A explicitly shown. The controller 20A features threesensor inputs: an intermediate-voltage input for an intermediate voltageV_(X), an output-voltage input for the output voltage V_(O), and anoptional input-voltage input for the input voltage V_(IN). Thecontroller 20A has two other inputs: a clock input to receive a clocksignal CLK and a reference input to receive a reference voltage V_(REF).Examples of the various signals above, as well as others to be describedbelow, can be seen in FIG. 14.

Based on the aforementioned inputs, the controller 20A provides a firstcontrol signal φ to control switches in the switched-capacitor element12A and a second control signal PWM to control switching of theregulating circuit 16A. The first control signal is a two-dimensionalvector having first and second complementary phases φ, φ. In someembodiments, the first control signal is a vector having higherdimensionality. In the illustrated embodiment, the second control signalPWM is a scalar. However, in multi-phase embodiments described below,the second control signal PWM is also a vector.

The controller 20A relies on the clock signal CLK and the intermediatevoltage V_(X) to set the period of the second control signal PWM forcontrolling the regulating circuit 16A. A comparison between thereference voltage V_(REF) and the output voltage V_(O) provides a basisfor controlling the output voltage V_(O).

The controller 20A synchronizes operation of the switching network 12Aand the regulating circuit 16A. It does so by synchronizing a ripple onthe intermediate voltage V_(X) with the second control signal PWM. Suchsynchronization relaxes the requirement of running the regulationcircuit 16A at a significantly higher frequency than the switchingnetwork 12A in an attempt to achieve effective feed-forward control.

The control method described herein also avoids glitches inherent inchanging the switching frequency of the switching network 12A. It doesso by making use of a regulating circuit 16A that draws discontinuousinput current. An example of such a regulating circuit 16A is one thatuses a buck converter.

Referring now to FIG. 13, the controller 20A has a switched-capacitorsection 301 and a regulator section 302.

The switched-capacitor section 301 outputs the first control signal φ.The complementary first and second phases φ, φ that make up the firstcontrol signal are shown as the last two traces in FIG. 14.

The switched-capacitor section 301 has an undershoot limiter 36 thatreceives the input voltage V_(IN) and the intermediate voltage V_(X).Based on these, the undershoot limiter 36 determines a trigger levelV_(X_L). The trigger level V_(X_L) is shown as a dashed horizontal linesuperimposed on the sixth trace on FIG. 14. The switched capacitorsection 301 ultimately uses this trigger level V_(X_L) to determine whenit is time to generate the first control signal φ. The details of howthis is done are described below.

After having generated the trigger level V_(X_L) based on the inputvoltage V_(IN) and the intermediate voltage V_(X), the undershootlimiter 36 provides it to a first comparator 35. The first comparator 35then compares the trigger level V_(X_L) with the intermediate signalV_(X). Based on the comparison, the first comparator 35 provides a firsttrigger signal to a first control signal generator 34, which ultimatelyoutputs the first control signal φ.

The switched capacitor section 301 thus forms a first feedback loop thatmanipulates the first control signal φ in an effort to control theintermediate voltage V_(X) based on the combination of the intermediatevoltage V_(X) and the input voltage V_(IN).

The first control signal generator 34 does not generate the firstcontrol signal φ immediately. Instead, the first control signalgenerator 34 waits for an opportune moment to do so. The occurrence ofthis opportune moment depends on what the regulator section 302 isdoing.

While the switched capacitor section 301 is busy providing the firsttrigger signal to the first control signal generator 34, the regulatorsection 302 is also busy generating the second control signal PWM. Theregulator section 302 begins this process with a voltage compensator 31that receives a voltage output V_(O) and a reference voltage V_(REF).From these, the voltage compensator 31 generates an error voltageV_(EER).

Some implementations of the voltage compensator 31 include linearvoltage-mode control and peak current-mode control. However, other modesare possible. Assuming linear voltage-mode control for the regulationcircuit 16A, the voltage compensator 31 compares the output voltageV_(O) of the power converter 10 with a reference voltage V_(REF) andprovides an error signal V_(ERR) to a second comparator 32. This errorsignal V_(ERR) is shown in FIG. 14 superimposed on a serrated waveformV_(SAW) on the second trace shown in FIG. 14.

The regulator section 302 thus forms a second feedback loop thatmanipulates the second control signal PWM in an effort to control theoutput voltage V_(O) based on the combination of a reference signalV_(REF) and the output voltage V_(O). However, for reasons discussed inmore detail below, the switched capacitor section 301 and the regulatorsection 302 do not operate independently. Instead, the controller 20Asynchronizes their operation.

To provide a basis for such synchronization, the regulator section 302includes a saw-tooth generator 30. The saw-tooth generator 30 generatesthe serrated waveform V_(SAW) based on a clock signal CLK and theintermediate voltage V_(X). This serrated waveform V_(SAW) ultimatelyprovides a way to synchronize the first control signal φ and the secondcontrol signal PWM.

The second comparator 32 compares the error voltage V_(ERR) with theserrated waveform VSAW and outputs a second trigger signal based on thiscomparison. As shown in FIG. 14, the second control signal PWM changesstate in response to a change in the sign of the difference between theerror voltage V_(ERR) and the serrated waveform V_(SAW). Since theserrated waveform V_(SAW) is ultimately based on the intermediatevoltage V_(X), this provides a basis for synchronizing the operation ofthe switched-capacitor section 301 and the regulator section 302.

The second control signal generator 33 receives the second triggersignal from the second comparator 32 and uses it as a basis forgenerating the second control signal PWM.

This second control signal PWM ultimately serves as a gate drive toactually drive the gate of a transistor that implements a main switch 52in a regulating circuit 16A, details of which are seen in FIG. 16. Thismain switch 52 ultimately controls an inductor voltage V_(L) and aninductor current I_(L) across and through an inductor 54 within theregulating circuit 16A, as shown by the fourth and fifth traces in FIG.14.

The particular configuration shown illustrates feed-forward control ofthe regulation circuit 16A implemented in the saw-tooth generator 30.However, such control could also be implemented in the voltagecompensator 31.

The switched-capacitor section 301 implements a hysteretic controlsystem in which a controlled variable, namely the intermediate voltageV_(X), switches abruptly between two states based upon a hysteresisband. The intermediate voltage V_(X) is a piecewise linear approximationof a serrated waveform.

Synchronization between the regulator section 302 and the switchedcapacitor section 301 is important to enable the dead-time interval ofthe switching network 12A to occur when no current is being drawn by theregulating circuit 16A.

In a practical switching network 12A, the first control signal φ willactually cycle through three states, not just two. In the first state,the first control signal φ opens a first set of switches and closes asecond set of switches. In the second state, the first control signal φcloses the first set of switches and opens the second set of switches.

A practical difficulty that arises is that switches cannot open andclose instantly. Nor can they be guaranteed to operate simultaneously.Thus, the first control signal φ cycles through a third state, whichlasts for a dead-time interval DT. During this third state, all switchesopen. This minimizes the unpleasant possibility that a switch in thesecond set will not have opened by the time the switches in the firstset have closed.

Meanwhile, certain regulating circuits 16A, such as buck converters andthe like, draw input current discontinuously. In particular, suchregulating circuits 16A have short intervals during which they aredrawing zero current.

The controller 20A avoids glitches by synchronizing the operation of theswitching network 12A and the regulating circuit 16A such that theregulating circuit 16A draws zero current during the dead-time intervalDT.

A further benefit of such synchronization is the ability to causeswitches in the switching network 12A to change state when there is nocurrent flowing through them. This reduces commutation losses. Causingthe dead-time interval DT to occur when the regulating circuit 16A isnot drawing current, and causing switches in the switching network 12Ato only change state at the beginning and the end of the dead-timeinterval DT thus ensures zero-current switching, as shown in FIG. 15.

In operation, the regulator section 302 and the switched capacitorsection 301 cooperate to ensure that the length of one cycle of thefirst control signal φ will be equal to an integral number of cycles ofthe second control signal PWM. In FIG. 14, this constraint is metbecause the one cycle of the first control signal φ is equal to anintegral number of cycles of the second control signal PWM.

The first control signal generator 34 receives a first trigger signalfrom the first comparator 35 indicating that the intermediate voltageV_(X) has fallen below the trigger level V_(X_L). However, as alluded toabove, the first control signal generator 34 does not act immediately.Instead, it waits until there is an opportune time to make a statechange. Meanwhile, as the first control signal generator 34 waits, theintermediate voltage V_(X) continues to fall, as shown in FIG. 14.

As shown in FIG. 14, by the time the first control signal generator 34acts, the intermediate voltage will already have fallen to an undershootΔV_(d) below the trigger level V_(X_L). In most cases, the undershootΔV_(d) is small and capped by an undershoot cap of ½ΔV_(X), which onlyoccurs when the switching frequency of the regulator section 302 and theswitched capacitor section 301 are equal. This undershoot cap depends onload current and input voltage V_(IN).

Large variations in undershoot ΔV_(d) are undesirable because theystress the regulating circuit 18A. The undershoot limiter 36 selects asuitable trigger level V_(X_L) to limit this undershoot ΔV_(d) byindirectly controlling the undershoot cap ½ΔV_(X). The undershootlimiter 36 uses the intermediate voltage V_(X) and the input voltageV_(IN) to select an appropriate value of the trigger level V_(X_L).

FIG. 15 shows a close up of selected waveforms in FIG. 14 at a scalethat is actually large enough to show a dead-time interval DT betweenthe two phases φ, φ that make up the first control signal φ. To assistin discussion, it is useful to consider the circuit shown in FIG. 16,which was introduced earlier in a discussion of the function of thesecond control signal PWM.

FIG. 16 shows a first set of switches 41, 43, 46, 48, which iscontrolled by the first phase φ, and a second set of switches 42, 44,45, 47, which is controlled by the second phase φ. FIG. 16 also showsthe main switch 52 that connects the regulating circuit 16A to theswitching network 12A. The main switch 52 has already been discussedabove.

During this dead-time interval DT, the phases φ, φ open all switches 41,43, 46, 48, 42, 44, 45, 47. This dead-time interval DT must occur whilethe main switch 52 is open. This requirement sets a maximum possibleduty cycle D_(max) for the regulating circuit 16A during the switchingtransition of the first control signal φ:

$D_{m\;{ax}} = \frac{T_{sw} - {DT}}{T_{sw}}$

As is apparent from the above relationship, the dead-time DT places alimit on the maximum possible duty cycle D_(max). It is thereforedesirable to reduce the dead-time DT as much as possible to increase therange of possible transformation ratios for the regulating circuit 16A.

For many practical power converters, a desire for electromagneticcompatibility dictates that the regulating circuit 16A should operate ata constant switching frequency. In these cases, the above constraint onthe maximum possible duty cycle D_(max) is not overly burdensome,especially, if the feed-back controller for the regulation circuit 16Awould otherwise have a maximum duty cycle requirement.

The control strategy as described above and implemented by thecontroller 20A in FIG. 13 is one of many possible implementations. Ingeneral, the switching frequency for switches 41, 43, 46, 48, 42, 44,45, 47 in the switching network 12A will change in discrete steps as theload current of the power converter 10 varies.

FIG. 17 shows how the output current affects both the period with whichthe switches 41, 43, 46, 48, 42, 44, 45, 47 of the switching network 12Achange state and the corresponding ΔV_(X) ripple.

For this particular control strategy, the ripple magnitude ΔV_(X) variesas a function of load current. In particular, the ripple magnitudeΔV_(X) defines a serrated waveform having a peak-to-peak amplitude thatdecreases with load current. As the load current approaches zero, thepeak-to-peak amplitude approaches half of the maximum peak-to-peakamplitude. With a few modifications to the controller, it is alsopossible to get the ΔV_(X) ripple to approach the maximum peak-to-peakamplitude as the load current approaches zero, as shown in FIG. 18.

As is apparent from both FIGS. 17 and 18, as the load current increases,the switching period for the switches 41, 43, 46, 48, 42, 44, 45, 47stays the same for a range of output currents. Within this range ofoutput currents, the converter relies on the regulating circuit 16A tomake up the difference between the voltage that the switching network12A provides whatever voltage is required. At some point, the regulatingcircuit 16A can no longer make the necessary correction. At that point,the period takes a step down.

The controller 20A shown in FIG. 12 is a single-phase converter. Assuch, the first control signal φ is a two-dimensional vector and thesecond control signal PWM is a scalar. In the case of an N-phaseconverter, the first control signal φ is a 2N-dimensional vector and thesecond control signal PWM is an N-dimensional vector having componentsPWM₁, PWM₂, . . . PWM_(n) that are phase shifted relative to each other.Typically, the phase shift between these components is 360/N degrees.

FIG. 19 shows an example of an N-phase converter having pluralregulation circuits 16A, 16B. Each regulation circuit 16A, 16B has acorresponding switching network 12A, 12B. Each regulation circuit 16A,16B is also driven by its own control signal, hence the need for anN-dimensional second control signal PWM. Each switching network 12A, 12Bis driven by a pair of phases, hence the need for a 2N-dimensional firstcontrol signal.

An N-phase controller 20A controls the N-phase converter. The N-phasecontroller 20A is similar to the single-phase controller in FIG. 12 butwith additional inputs for the N intermediate voltages V_(X1), V_(X2), .. . V_(XN).

FIG. 20 shows waveforms similar to those shown in FIG. 14 but for athree-phase version of the controller shown in FIG. 12.

As shown in FIG. 20, the second control signal PWM consists of secondcontrol signal elements PWM₁, PWM₂, PWM₃ that are separated from eachother by a delay time that corresponds to a 120° phase shift betweenthem. The three intermediate voltages V_(X1), V_(X2), V_(X3) are shiftedfrom each other by an integer multiple of this delay time. In FIG. 20,the integer is unity. However, as shown in FIG. 21, other integers arepossible.

Because the periods of the intermediate voltages V_(X1), V_(X2), V_(X3)are longer than those of the second control signal elements PWM₁, PWM₂,PWM₃, shifting them by the delay time will not cause them to be 120degrees out of phase with each other. In fact, because their period isso much longer, a shift by this delay time only causes a very smallphase shift in the intermediate voltages V_(X1), V_(X2), V_(X3).

FIG. 21 shows an alternative method of operation similar to that shownin FIG. 20, but with the intermediate voltages V_(X1), V_(X2), V_(X3)having been shifted by a larger multiple of the delay time. This resultsin a more significant phase shift between the intermediate voltagesV_(X1), V_(X2), V_(X3), a result of which is a reduced ripple in theoutput voltage V_(O).

A multi-phase controller 20A for controlling the N-phase converter shownin FIG. 19 can be thought of as N single phase controllers 20A as shownin FIG. 13 operating in parallel but with a specific phase relationshipbetween them. A multi-phase controller 20A would thus look very similarto the one in FIG. 13, but with an additional input and output signals.In general, the intermediate voltages (V_(X1), V_(X2), V_(XN)) and theoutput voltage V_(O) are required for proper operation of the controller20A.

In FIG. 12, a non-capacitive regulating circuit 16A loads down theswitching network 12A. This regulating circuit 16A is switched at a highfrequency. The components from the high-frequency switching of theregulating circuit 16A are ultimately superimposed on the lowerfrequency serrated waveform of the intermediate voltage V_(X), as shownin sixth trace on FIG. 14. The duty cycle of the saw-tooth approximationwaveform depends on the topology of the switching network 12A. Ingeneral, the frequency of the complementary switching-network controlsignals varies with changes in response to changes in the slope of theintermediate signal. These changes, in turn, arise as a result ofchanges in the power converter's operating point.

The switching network 12A and the regulating circuit 16A are essentiallymodular and can be mixed and matched in a variety of different ways. Assuch, the configuration shown in FIG. 11 represents only one of multipleways to configure one or more switching networks 12A with one or moreregulating circuits 16A to form a multi-stage DC-DC converter 10 of apower converter.

For example, FIG. 22 shows a bidirectional version of FIG.11 in whichpower can flow either from a voltage source 14 to a load 18A or from theload 18A to the voltage source 14 as indicated by the arrows.

There are two fundamental elements described in connection with thefollowing embodiments: switching networks 12A and regulating circuits16A. Assuming series connected elements of the same type are combined,there are a total of four basic building blocks. These are shown FIGS.22, 23, 24, and 26. The power converters disclosed herein include atleast one of the four basic building blocks. More complex converter canbe realized by combining the fundamental building blocks.

The first building block, shown in FIG. 22, features a switching network12A whose output connects to an input of a regulating circuit 16A. Thesecond building block, shown in FIG. 23, features a first switchingnetwork 12A whose output connects to a regulating circuit 16A, an outputof which connects to an input of a second switching network 12B. In thethird building block shown in FIG. 24, an output of a regulating circuit16A connects to an input of a switching network 12A. A fourth buildingblock, shown in FIG. 27, features a first regulating circuit 300A havingan output that connects to an input of a first switching network 200, anoutput of which connects to an input of a second regulating circuit300B.

Additional embodiments further contemplate the application ofobject-oriented programming concepts to the design of power convertersby enabling switching networks 12A and regulating circuits 16A to be“instantiated” in a variety of different ways so long as their inputsand outputs continue to match in a way that facilitates modular assemblyof power converters having various properties.

The switching network 12A in many embodiments is instantiated as aswitched-capacitor network. Among the more useful switched capacitortopologies are: Ladder, Dickson, Series-Parallel, Fibonacci, andDoubler, all of which can be adiabatically charged and configured intomulti-phase networks. A particularly useful switching capacitor networkis an adiabatically charged version of a full-wave cascade multiplier.However, diabatically charged versions can also be used.

As used herein, changing the charge on a capacitor “adiabatically” meanscausing an amount of charge stored in that capacitor to change bypassing the charge through a non-capacitive element. A positiveadiabatic change in charge on the capacitor is considered adiabaticcharging while a negative adiabatic change in charge on the capacitor isconsidered adiabatic discharging. Examples of non-capacitive elementsinclude inductors, magnetic elements, resistors, and combinationsthereof.

In some cases, a capacitor can be charged adiabatically for part of thetime and diabatically for the rest of the time. Such capacitors areconsidered to be adiabatically charged. Similarly, in some cases, acapacitor can be discharged adiabatically for part of the time anddiabatically for the rest of the time. Such capacitors are considered tobe adiabatically discharged.

Diabatic charging includes all charging that is not adiabatic anddiabatic discharging includes all discharging that is not adiabatic.

As used herein, an “adiabatically charged switching network” is aswitching network having at least one capacitor that is bothadiabatically charged and adiabatically discharged. A “diabaticallycharged switching network” is a switching network that is not anadiabatically charged switching network.

The regulating circuit 16A can be instantiated as any converter with theability to regulate the output voltage. A buck converter for example, isan attractive candidate due to its high efficiency and speed. Othersuitable regulating circuits 16A include boost converters, buck/boostconverters, fly-back converters, forward converters, half-bridgeconverters, full-bridge converters, Cuk converters, resonant converters,and linear regulators. The fly-back converter can more specifically be aquasi-resonant fly-back converter, or an active-clamp fly-backconverter, or an interleaved fly-back converter, or a two-switchfly-back converter. Likewise, the forward converter can be morespecifically a multi-resonant forward converter, or an active-clampforward converter, or an interleaved forward converter, or a two-switchforward converter. And, the half-bridge converter can more specificallybe an asymmetric half-bridge converter, or a multi-resonant half-bridgeconverter, or a LLC resonant half-bridge.

In the embodiment shown in FIG. 22, a source voltage 14 provides aninput to a first switching network 12A, which is instantiated as aswitching capacitor network. The output of the first switching network12A is a lower voltage than the input voltage that is provided to aregulating circuit 16A (e.g. a buck, a boost, or a buck/boostconverter). This regulating circuit 16A provides a regulated inputvoltage to a second switching network 12B, such as another switchingcapacitor network. A high voltage output of this second switchingnetwork 12B is then applied to a load 18A.

An embodiment such as that shown in FIG. 22 can be configured toregulate the load 18A or to regulate the voltage source 14 depending onthe direction of energy flow.

In another embodiment, shown in FIG. 24, a low voltage source 14connects to an input of a regulating circuit 16A, the output of which isprovided to an input of a switching network 12A to be boosted to ahigher DC value. The output of the switching network is then provided toa load 18A.

An embodiment such as that shown in FIG. 24 can be used to regulate thevoltage source 14 or the load 18A depending on the direction of energyflow.

FIG. 25 shows the modular DC-DC converter 10C of FIG. 24, but with acontroller 20A explicitly shown. The controller 20A is similar to thatdescribed in connection with FIG. 13.

As was discussed in connection with FIG. 13, the controller 20A featuresthree sensor inputs, one for an intermediate voltage V_(X), one for theoutput voltage V_(O), and an optional one for the input voltage, V_(IN).The controller 20A also has two inputs that are not sensor inputs. Onenon-sensor input receives a clock signal CLK and the other receives areference voltage V_(REF). The clock signal CLK is used to set theperiod of a second control signal PWM and the reference voltage V_(REF)is used to set the desired output voltage. Based on these inputs, thecontroller 20A outputs a first control signal having two phases to theswitched-capacitor element 12A and a second control signal PWM tocontrol switching of the regulating circuit 16A. This second controlsignal PWM is a pulse-width modulated signal.

Referring now to FIG. 26, another embodiment of a converter 100 includesa first regulating circuit 300A connected to a converter input 102 and asecond regulating circuit 300B connected to a converter output 104.Between the first and second regulating circuits 300A, 300B is aswitching network 200 having a switching network input 202 and aswitching network output 204. The switching network 200 includes chargestorage elements 210 interconnected by switches 212. These chargestorage elements 210 are divided into first and second groups 206, 208.

In some embodiments, the switching network 200 is a bidirectionalswitching capacitor network such as that shown in FIG. 27.

The switching capacitor network in FIG. 27 features a first capacitor 20and a second capacitor 22 in parallel. A first switch 24 selectivelyconnects one of the first and second capacitors 20, 22 to a firstregulating circuit 300A, and a second switch 26 selectively connects oneof the first and second capacitors 20, 22 to the second regulatingcircuit 300B. Both the first and second switches 24, 26 can be operatedat high frequency, thus facilitating the adiabatic charging anddischarging of the first and second capacitors 20, 22.

The particular embodiment shown in FIG. 27 has a two-phase switchingnetwork 200. However, other types of switching networks can be usedinstead.

In yet another embodiment, shown in FIG. 28, multiple regulatingcircuits 16A, 16B, 16C are provided at an output of a first switchingnetwork 12A for driving multiple loads 18A-18C. For one of the loads18C, a second switching network 12B is provided between the load 18C andthe corresponding regulating circuit 16C thus creating a pathway similarto that shown in FIG. 24. FIG. 28 thus provides an example of how themodular construction of regulating circuits and switching networksfacilitates the ability to mix and match components to provideflexibility in DC-DC converter construction.

A switched-capacitor power converter includes a network of switches andcapacitors. By cycling the network through different topological statesusing these switches, one can transfer energy from an input to an outputof the switched-capacitor network. Some converters, known as “chargepumps,” can be used to produce high voltages in FLASH and otherreprogrammable memories.

To help understand the loss mechanisms in switched capacitor converters,it is instructive to first analyze the classical capacitor chargingproblem, as depicted in FIG. 29.

FIG. 29 shows a capacitor C initially charged to some value V_(C)(0). Att=0 the switch S is closed. At that instant, a brief surge of currentflows as the capacitor C charges to its final value of V_(IN). The rateof charging can be described by a time constant τ=RC, which indicatesthe time it takes the voltage to either rise or fall to within 1/e ofits final value. The instantaneous values for voltage across thecapacitor v_(c)(t) and current through the capacitor i_(c)(t) are givenby the following equations:

v_(c)(t) = v_(c)(0) + [V_(i n) − v_(c)(0)](1 − e^(−t/RC)), and${i_{c}(t)} = {{C\;\frac{{dv}_{c}}{dt}} = {\frac{V_{i\; n} - {v_{c}(0)}}{R}{e^{{- t}/{RC}}.}}}$

The energy loss incurred while charging the capacitor can be found bycalculating the energy dissipated by resistor R, which isE _(loss)(t)=∫_(t=0) ^(∞) i _(R)(t)×v _(R)(t)dt=∫ _(t=0) ^(∞) [i_(c)(t)]² R dt.

The equation can be further simplified by substituting the expressionfor i_(c)(t) into the equation above. Evaluating the integral thenyieldsE _(loss)(t)=1/2[V _(in) −v _(c)(0)]² C[1−e ^(−2t/RC)].

It is apparent therefore that the only term that involves the resistanceis in a decaying exponential. Thus, if the transients are allowed tosettle (i.e. t→∞), the total energy loss incurred in charging thecapacitor is independent of its resistance R. In that case, the amountof energy loss is equal toE _(loss)(∞)=1/2CΔv _(c) ².

A switched-capacitor converter can be modeled as an ideal transformer,as shown in FIG. 30, with a finite output resistance R_(o) that accountsfor the power loss incurred in charging or discharging of the energytransfer capacitors, as shown in FIG. 30. The embodiment shown in FIG.30 is non-isolated because the negative terminals on both sides of thetransformer are connected. However, this is by no means required. As anexample, FIG. 31 shows an embodiment in which the same terminals are notconnected, in which case the converter is isolated.

It should be noted that the transformer shown is only for modelingpurpose. A converter of this type would generally not have windingswrapped around an iron core. The power losses associated with chargingand discharging are typically dissipated in the ON resistance of theMOSFETs and equivalent series resistance of the capacitors.

The output voltage of the switched-capacitor converter is given by

${V_{o} = {{V_{in}\frac{N_{2}}{N_{1}}} - {I_{o}R_{o}}}}.$

There are two limiting cases where the operation of switched capacitorconverters can be simplified and R_(o) easily found. These are referredto as the “slow-switching limit” and the “fast-switching limit.”

In the fast-switching limit (τ>>T_(sw)), the charging and dischargingcurrents are approximately constant, resulting in a triangular AC rippleon the capacitors. Hence, R_(o) is sensitive to the series resistance ofthe MOSFETs and capacitors, but is not a function of the operatingfrequency. In this case, R_(o) of the converter operating in thefast-switching limit is a function of parasitic resistance and R_(o) isgiven by:

${\left. R_{o} \right|_{\tau\bullet T_{sw}} = {R_{FSL} = {n{\sum\limits_{i \in {sw}}{\sum\limits_{j = 1}^{n}{R_{i}\left( a_{r,i}^{j} \right)}^{2}}}}}}.$

Although it tends to under-estimate R_(o), a useful approximation forR_(o) that serves as a good starting point in the design process isgiven byR _(o)(f)≈√{square root over (R _(FSL) ² +R _(SSL) ²)}.

In the slow-switching limit, the switching period T_(sw) is much longerthan the RC time constant τ of the energy transfer capacitors. Underthis condition, a systemic energy loss given by 1/2C×ΔV_(c) ² occursregardless of the resistances of the capacitors and switches. Thissystemic energy loss arises in part because the root mean square (RMS)of the charging and discharging current is a function of the RC timeconstant. Under these circumstances, R_(o) is given by

$\left. R_{o} \right|_{\tau\bullet T_{sw}} = {R_{SSL} = {\sum\limits_{i \in {caps}}{\sum\limits_{j = 1}^{n}{\frac{\left( a_{c,i}^{j} \right)^{2}}{2C_{i}f_{sw}}.}}}}$

The behavior of output resistance as a function of frequency can beappreciated by inspection of FIG. 32, which shows that as frequencyincreases, the output resistance drops in a manner consistent with the1/f_(sw) term and that at higher frequencies, the output resistancesettles down to a steady value.

The calculations for R_(SSL) and R_(FSL) given above are based on thecharge multiplier vector concept. The vector a¹ through a^(n) can beobtained by inspection for any standard well posed n-phase converter.The charge multiplier vectors are computed using constraints imposed byKirchoff's current law in each topological state along with thesteady-state constraint that the n charge multiplier quantities must sumto zero on each capacitor.

Once R_(o) is known, the conduction loss P_(cond) can be calculated byP_(cond)=I_(o) ²R_(o).

Additionally, other losses such as switching losses, driver losses, andcontrol losses can be calculated. Preferably, the switching loss iscomparable to conduction loss. These losses, which originate fromcharging and discharging the transistor nodes, are given byP _(sw) =W _(sw) f _(sw)=(W _(ds) +W _(on) +W _(g))f _(sw)where W_(g) is the gate capacitance loss, W_(on) is the overlap orcommutation loss, and W_(ds) is the output capacitance loss. Thus thetotal converter loss can be calculated usingP _(loss) =I _(o) ² R _(o) +W _(sw) f _(sw) +P _(etc).Once R_(o) and the additional loss mechanisms have been determined, thetotal efficiency of the converter is given by

${\eta_{sc} = {\frac{P_{o}}{P_{o} + P_{loss}} = \frac{P_{o}}{P_{o} + P_{cond} + P_{sw} + P_{etc}}}}.$

To optimize efficiency of the switched-capacitor converter, the optimalswitching frequency, capacitance, and device sizes must be selected. Ifthe switching frequency is too low, then the conduction losses,P_(cond), dominate. On the other hand, if the switching frequency is toohigh, then P_(sw) dominates. Although doing so tends to decrease outputripple, rarely will a switched-capacitor converter operate far above thetransitional region between the slow switching limit and fast switchinglimit. After all, operating above this region tends to increaseswitching losses without lowering the output resistance to compensatefor those increases switching losses. Thus, there is little to gain byoperating above that region.

If the effective resistance R_(eff) of the charging path is reduced, forexample by reducing the RC time constant, the RMS current increases andit so happens that the total charging energy loss (E_(loss)=I_(RMS)²R_(eff)=1/2C×ΔV_(C2)) is independent of R_(eff). One solution tominimize this energy loss is to increase the size of the pump capacitorsin the switched capacitor network.

Although many switched-capacitor networks can provide a specific voltagetransformation, most of them are impractical for a variety of reasons. Apractical switched-capacitor network typically has a largetransformation ratio, low switch stress, low DC capacitor voltage, andlow output resistance. Suitable topologies for the converters describedherein include Ladder, Dickson, Series-Parallel, Fibonacci, and Doublertopologies.

One useful converter is a series-parallel switched capacitor converter.FIGS. 33-34 show a 2:1 series-parallel switched-capacitor converteroperating in charge phase and in discharge phase respectively. Duringthe charge phase, the capacitors are in series. In the discharge phase,the capacitors are in parallel. In its charge phase the capacitorvoltages v_(c1) and v_(c2) add up to V₁ while in its discharge phasev_(c1) and v_(c2) equal V₂. This means that V₂=V_(1/2).

Another useful family of switched-capacitor topologies is that firstdiscovered by Greinacher and popularized by Cockcroft, Walton, andDickson. An example of such a topology is that shown in FIGS. 35 and 36.In both charge pumps, the source is located at V₁ and the load islocated at V₂. In these types of charge pumps, packets of charge arepumped along a diode chain as the coupling capacitors are successivelycharged and discharged. As shown in FIG. 37, clock signals v_(clk) andv_(clk) with amplitude v_(pump) are 180 degrees out of phase. Thecoupling capacitors can either be pumped in series or parallel.

It takes n clock cycles for the initial charge to reach the output. Thecharge on the final pump capacitor is n times larger than the charge onthe initial pump capacitor. Thus, V₂ for the converters in FIG. 36 isV₁+(n−1)×v_(pump) in both pumping configurations.

Although the foregoing topologies are suitable for stepping up voltage,they can also be used to step down voltage by switching the location ofthe source and the load. In such cases, the diodes can be replaced withcontrolled switches such as MOSFETs and BJTs.

FIGS. 35 and 36 show topologies that transfer charge during only onephase of the clock signal. Such topologies are referred to as“half-wave” topologies because charge transfer only occurs during halfof a clock cycle. A disadvantage of a half-wave topology is adiscontinuous input current.

It is possible to convert the topologies shown in FIGS. 35 and 36 sothat they transfer charge during both phases of the clock signal. Thiscan be carried out by connecting two such topologies in parallel anddriving them 180 degrees out of phase. Such a topology is referred toherein as a “full-wave” topology because charge transfer occurs in bothhalves of the clock cycle.

FIG. 38 show a topology derived from that shown in FIG. 35, but modifiedso that charge transfer occurs in both phases of the clock signal. FIG.39 show a topology derived from that shown in FIG.36, but modified sothat charge transfer occurs in both phases of the clock signals. Insteadof diodes, as shown in the topologies of FIGS. 35 and 36, the topologiesshown in FIGS. 38 and 39 use switches. Unlike diodes, which areinherently unidirectional, the switches shown in FIG. 38 and FIG. 39 arebidirectional. As a result, in the topologies shown in FIGS. 38 and 39,power can flow either from the V₁ terminal to the V₂ terminal or viceversa. As such, these topologies can be used to step-up a voltage orstep-down a voltage.

In the topologies shown thus far, there are two chains of switches, eachof which is pumped. However, it is also possible to pump only one of thetwo switch chains. Such topologies are referred to as “asymmetric.”

In asymmetric topologies, half of the capacitors are used to support aDC voltage and not to transfer energy. However, these embodiments do notrequire that each switch endure such a high peak voltage. In particular,the peak voltage in the case in which only one switch chain is beingpumped is only half of what it would be if both switch chains wereactually being pumped. In these asymmetric topologies, the sole switchchain that is being used to transfer energy can be modified to transfercharge during both phases of the clock signal using principles set forthin connection with FIG. 38.

FIG. 40 shows eight exemplary topologies that use the principles setforth in connection with FIGS. 35-39. The first and second columns showhalf-wave topologies in both asymmetric and symmetric configurations,whereas the third and fourth columns show full-wave wave topologies inboth asymmetric and symmetric configurations. The topologies shown inFIG. 40 can be further modified to combine N phases in parallel and torun them 180 degrees/N out of phase. Doing so reduces output voltageripple and increases output power handling capability.

The basic building blocks in the modular architecture shown FIGS. 22,23, 24, and 26 can either be connected as independent entities orcoupled entities. In the situation where switching networks andregulating circuits are tightly coupled, it is possible to preventand/or reduce the systemic energy loss mechanism of the switchingnetworks through adiabatic charging. This generally includes using aregulating circuit to control the charging and discharging of thecapacitors in the switching network. Furthermore, the output voltage ofthe regulating circuit and thus the total converter can be regulated inresponse to external stimuli. One approach to regulating the outputvoltage is by controlling the average DC current in the magnetic storageelement.

In general, it is desirable for the regulating circuit to operate in away that limits the root mean square (RMS) current through thecapacitors in the switching network. The regulating circuit can do sousing either resistive elements or magnetic storage elements. Becauseresistive elements consume power, magnetic storage elements aregenerally preferable for this purpose. Therefore, embodiments describedherein rely on a combination of switches and a magnetic storage elementin the regulating circuit to limit RMS current in the switching network.

To limit RMS current, the regulating circuit forces the capacitorcurrent through the magnetic storage element in a regulating circuitthat has an average DC current. The regulating circuit's switches thenoperate so as to maintain an average DC current through the magneticstorage element.

The regulating circuit may limit both the RMS charging current and theRMS discharging current of at least one capacitor in the switchingnetwork. A single regulating circuit may limit the current into or outof the switching network by sinking and/or sourcing current. Therefore,there are four fundamental configurations, which are shown in FIGS. 22,23, 24, and 26.

Assuming power flows from source to load then, in FIG. 22, theregulating circuit 16A may sink both the charging and dischargingcurrent of the switching network 12A.

In FIG. 23, the regulating circuit 16A may source both the charging anddischarging current of the switching network 12B while also sinking boththe charging and discharging current of the switching network 12A.Furthermore, if both the switching networks and the regulating circuitsallow power to flow in both directions, then bidirectional power flow ispossible.

In FIG. 24, the regulating circuit 16A may source both the charging anddischarging current of the switching network 12A.

In FIG. 26, the regulating circuit 300A may source the charging currentof switching network 200 and the regulating circuit 300B may sink thedischarging current of the same switching network 200 and vice-versa.

A fundamental difficulty that afflicts switched-capacitor networks isthat the mere act of charging a capacitor incurs energy loss. Thisenergy loss depends a great deal on how much the voltage across thecapacitor changes as a result of the charging event. The energy lossE_(L) associated with using a fixed voltage source at a voltage V tocharge a capacitance C from zero to V is 1/2CV². This loss does notdepend on the parasitic series resistance R. Since this loss ariseswhenever voltage changes, every charging interval during operationincurs a loss equal to 1/2CΔV², where ΔV corresponds to the differencebetween the initial and final value of the capacitor voltage.

The fixed charge-up loss cannot be reduced by employing switches withlower on-state resistance. Known ways to reduce it simply avoid causingthe voltage to change very much during operation. This is why suchconverters operate most efficiently only at certain conversion ratios.

Since the amount of charge transferred into or out of a charging cycleis the product of the voltage difference and the capacitance, one way totransfer a great deal of charge with only a small voltage difference isto make the capacitance very large. However, large capacitors are notwithout disadvantages. For one thing, a large capacitance consumes agreat deal of physical area. Additionally, switched-capacitor networkswith large capacitances are not so amenable to efficient operation.

A converter as described herein overcomes the foregoing disadvantage byproviding more efficient use of the capacitors. This means thatcapacitors can be made smaller and/or that there will be an overallimprovement in system efficiency. Although a converter as describedherein does not require a reconfigurable switched-capacitor circuit, itmay nevertheless take advantage of one as described above.

FIG. 41 illustrates a method for improving the charge-up efficiency ofthe capacitor C shown in FIG. 29 after switch S closes. The regulatingcircuit 16A adiabatically charges the capacitor C. In some embodiments,the regulating circuit 16A is a switch-mode converter that supplies anoutput. A suitable regulating circuit is a low-voltage magnetic basedconverter.

In the system shown in FIG. 41, while the capacitor C charges, most ofthe difference between the input voltage V_(IN) and the capacitor stackvoltage V_(C) appears across the input of the regulating circuit 16A.Instead of being dissipated as heat in a parasitic resistor R, theenergy associated with charging the capacitor stack is delivered to theoutput of the regulating circuit 16A instead. Therefore, a majority ofthe capacitor-charging energy can be recovered (i.e., redirected to theload) by making the apparent input resistance of the regulating circuit16A higher than the parasitic resistor R.

The embodiment shown in FIG. 41 thus permits more efficient use ofcapacitors than that shown in FIG. 29. This enables reduction in therequired capacitor size and/or improvement in system efficiency whenextended to switched-capacitor converters.

FIG. 42 illustrates one implementation of the foregoing embodiment inwhich a switching network 12A connects to regulating circuit 16A thatserves as both a means to adiabatically charge/discharge the capacitorsin the switching network 12A and regulate the output voltage V_(O).Please note, the regulating circuit 16A need not be at a higherfrequency than the switching network to promote adiabatic operation; itcan even be at a lower frequency. In the particular embodiment shown,the regulating circuit 16A is a synchronous buck converter and theswitching network 12A is a single-phase series-parallel converter. Theswitching network 12A features first switches 1 that open and closetogether, second switches 2 that also open and close together, a firstpump capacitor C₁, and a second pump capacitor C₂.

The regulating circuit 16A includes a filter capacitor C_(X) that servesonly as a filter and bypass for the regulating circuit 16A.Consequently, the capacitance of the filter capacitor C_(X) should bemuch smaller than that of the first and second pump capacitors C₁ and C₂of the switching network 12A.

The switching network 12A alternates between being in a charging stateand a discharging state. During the charging state, it charges the firstand second pump capacitors C₁, C₂. Then, during the discharging state,it discharges the first and second pump capacitors C₁, C₂ in parallel.

In the charging state, the first switches 1 close and the secondswitches 2 open. The difference between the input voltage V_(IN), andthe sum of the voltages across the first and second pump capacitors C₁,C₂ appears across the input terminal of the regulating circuit 16A. As aresult, the first and second pump capacitors C₁, C₂ charge with lowloss, and at a rate determined by the power drawn from the regulatingcircuit 16A to control the system output.

Similarly, in the discharging state, the second switches 2 close and thefirst switches 1 open. The switching network 12A then discharge inparallel at a rate based on the power needed to regulate the output.

Another embodiment relies on at least partially adiabatically chargingfull-wave cascade multipliers. Cascade multipliers are a preferredswitching network because of their superior fast-switching limitimpedance, ease of scaling up in voltage, their two phase operation, andlow switch stress.

In cascade multipliers, the coupling capacitors are typically pumpedwith a clocked voltage source v_(clk) & v_(clk) . However, if thecoupling capacitors are pumped with a clocked current source i_(clk) &i_(clk) instead, as shown in FIG. 43, then the RMS charging anddischarging current in the coupling capacitor may be limited. In thiscase, the capacitors are at least partially charged adiabatically thuslowering, if not eliminating, the 1/2CΔV_(c) ² loss that is associatedwith a switched-capacitor converter when operated in the slow-switchinglimit. This has the effect of lowering the output impedance to thefast-switching limit impedance. As shown by the black dotted line inFIG. 44, which depicts adiabatic operation under full adiabaticcharging, the output impedance would no longer be a function ofswitching frequency.

With all else being equal, an adiabatically charged switched-capacitorconverter can operate at a much lower switching frequency than aconventionally charged switched-capacitor converter, but at higherefficiency. Conversely, an adiabatically charged switched-capacitorconverter can operate at the same frequency and with the same efficiencyas a conventionally charged switched-capacitor converter, but with muchsmaller coupling capacitors, for example between four and ten timessmaller.

Embodiments described herein can operate with two clocked currentsources i_(clk), i_(clk) that operate 180 degrees out of phase, as shownin FIG. 45. One implementation, shown in FIG. 46, uses one currentsource 72, a first switch pair 1 and a second switch pair 2. The firstand second switch pairs 1, 2 are best synchronized with a switch chain.A suitable implementation of the current source in FIG. 46 is aninductance, represented in FIG. 47 by an inductor L.

FIG. 48 shows the cascade multiplier of FIG. 43 with the clocked currentsources in FIG. 46. There are numerous ways of implementing the currentsource 72. These include buck converters, boost converters, fly-backconverter, resonant converters, and linear regulators. In someembodiments, a power converter having a constant input currentimplements the constant current source. In other embodiments, a powerconverter that has a constant input current for a portion of an intervaldefined by the reciprocal of its switching frequency implements theconstant current source. In yet other embodiments, a linear regulatorimplements the constant current source.

FIG. 49 shows a step-down converter consistent with the architectureshown in FIG. 22. However, in this embodiment, a switching network 12Ais adiabatically charged using a regulating circuit 16A. The clockedcurrent sources i_(clk) & i_(clk) are emulated by four switches and theregulating circuit 16A. The output capacitor Co has also been removed soas to allow V_(X) to swing. In this example, the regulating circuit 16Ais a boost converter that behaves as constant source with a small ACripple. Any power converter that has a non-capacitive input impedance atthe frequency of operation would have allowed adiabatic operation.Although switch-mode power converters are attractive candidates due totheir high efficiency, linear regulators are also practical.

In operation, closing switches labeled “1” charges capacitors C₄, C₅,and C₆ while discharging capacitors C₁, C₂, and C₃. Similarly, closingswitches “2” has the complementary effect. The first topological state(phase A) is shown in FIG. 49, where all switches labeled “1” are closedand all switches labeled “2” are opened. Similarly, the secondtopological state (phase B) is shown in FIG. 50, where all switcheslabeled “2” are closed and all switches labeled “1” are opened.

In this embodiment, the regulating circuit 16A limits the RMS charge anddischarging current of each capacitor. For example, capacitor C₃ isdischarged through the filter inductor in the regulating circuit 16Aduring phase A, while capacitor C₃ is charged through the filterinductor in regulating circuit 16A during phase B, clearly demonstratingthe adiabatic concept. Furthermore, all of the active components areimplemented with switches so that the converter can process power inboth directions.

A few representative node voltages and currents are shown in FIG. 52.There is a slight amount of distortion on the rising and falling edgesof the two illustrated currents (I_(P1) and I_(P2)), but for the mostpart, the currents resemble two clocks 180 degrees out of phase. Ingeneral, adiabatic charging occurs in cascade multipliers if at leastone end of a switch stack is not loaded with a large capacitance, as isthe case in this embodiment, where the V_(X) node is loaded down byregulating circuit 16A.

In operation, different amounts of current will flow through differentswitches. It is therefore useful to size the switches in a mannerappropriate to the currents that will be flowing through them. Forexample, the switches connected to V_(P1) and V_(P2) carry more currentthen the other switches in FIG. 49. By making these switches larger thanthe other switches, this avoids the need to have unnecessarily largeswitches and thus results in a smaller circuit footprint. This alsoavoids unnecessary additional capacitive losses, which are proportionalto the size of the switch.

The switches shown in FIG. 49 will transition between states at someswitching frequency. It is desirable that, in order to reduce loss, theswitching network 12A operate such that the RMS current through theswitches is constrained at that switching frequency. One way to ensurethat this is the case is to choose the resistances of the switches suchthat they are so large that the RC time constant of the charge transferbetween the capacitors is similar if not longer than the switchingfrequency. As can be seen in FIG. 44, by controlling the width “W” ofthe switches and hence their resistance and their size, the switchingnetwork 12A can be forced into the fast-switching limit region.

Unfortunately, by using the resistance of the switches to constrain theRMS current, conductive power losses increase and the overall efficiencydecreases. The regulating circuit 16A, however, allows us to reduce theresistance of the switches and operate adiabatically. Therefore, theswitches can be optimally sized for the highest efficiency withoutworrying about constraining the RMS current since it is handled by theregulating circuit 16A (or optionally a magnetic filter). The optimalsize for each switch is chosen by balancing the resistive and capacitivelosses in each switch at a given switching frequency and at a givencurrent.

The modular architecture with the basic building blocks shown in FIGS.11, 23, 24, and 26 may be expanded to cover a wider range ofapplications, such as high-voltage DC, AC-DC, AC-AC, buck-boost, andmultiple output voltages. Each of these applications includes separatingthe transformation and regulation functions. Extension of thearchitecture can also incorporate adiabatically chargedswitched-capacitor converters.

In many switched-capacitor converters, the number of capacitors andswitches increases linearly with the transformation ratio. Thus, a largenumber of capacitors and switches are required if the transformationratio is large. Alternatively, a large transformation ratio can beachieved by connecting numerous low gain stages in series as depicted inFIG. 53. The transformation ratio of the total switch capacitor stack(V_(IN)/V_(X)) is as follows:

$\begin{matrix}{\frac{V_{in}}{V_{x}} = {N_{1} \times N_{2}\mspace{14mu}\ldots\mspace{14mu} N_{n}}} & (2.1)\end{matrix}$

The main disadvantage of the series stacked configuration is that thevoltage stresses on the front stages are much higher than those of therear stages. This will normally require stages with different voltageratings and sizes. However, the transformation ratio can be easilychanged by bypassing a stage or two.

Adiabatic charging of a preceding series-connected switching networkonly occurs if the following switching network controls the charging anddischarging current of the preceding stage. Thus, it is preferable touse full-wave switched-capacitor converters in the front stages or touse switched-capacitor stages such as the single-phase series-parallelswitched-capacitor converters with magnetic based filters.

FIG. 54 shows a converter with two series-connected switching networksconsistent with the architecture shown in FIG. 53. Both switchingnetworks 12A, 12D are two-phase cascade multipliers. In operation,switches labeled “1” and “2” are always in complementary states andswitches labeled “7” and “8” are always in complementary states. Thus,in a first switched-state, all switches labeled “1” are open and allswitches labeled “2” are closed. In a second switched-state, allswitches labeled “1” are closed and all switches labeled “2” are opened.In this embodiment, closing switches 1 charges capacitors C₁, C₂, C₃,while discharging capacitors C₄, C₅, C₆ and closing switches 2 has thecomplementary effect. Also, closing switches 7 charges capacitors C₇,C₈, C₉, while discharging capacitors C₁₀, C₁₁, C₁₂ and closing switches8 has the complementary effect.

The power converter provides a total step-down of 32:1, assuming theregulating circuit 16A is a buck converter with a nominal step-downratio of 2:1. Furthermore, if the input voltage is 32 V and the outputvoltage is 1 V, then the switches in the first switching network 12Awill need to block 8 volts while the switches in the second switchingnetwork 12D will need to block 2 volts.

The modular architecture with the basic building blocks shown in FIGS.11, 23, 24, and 26 may be configured to handle an AC input voltage asshown in FIG. 55. An AC rectification stage 19A receives an AC waveformfrom an AC source 14B and provides an average DC voltage to a modularDC-DC converter 10, the output of which is connected to a load 18A. Inthis embodiment, the modular DC-DC converter 10 can be isolated orotherwise.

One of the main attributes of switched-capacitor converters is theirability to operate efficiency over a large input range by reconfiguringthe switched-capacitor network. If the AC wall voltage (i.e. 60 Hz & 120V_(RMS)) can be thought of as a slow moving DC voltage, then a front-endAC switching network 13A should be able to unfold the time-varying inputvoltage into a relatively stable DC voltage.

FIG. 56 shows a diagram of a 120 V_(RMS) AC waveform over a single 60 Hzcycle overlaid with the unfolded DC voltage. FIG. 57 shows an ACswitching network 13A of the sort that can incorporate the ACrectification stage 19A of FIG. 55. The AC switching network 13A is afront-end switched-capacitor stage (i.e., switching network) incombination with a selective inverting stage (i.e., rectifying stage).The front-end switched-capacitor stage has different configurations(1/3, 1/2, 1/1) at its disposal. In the particular embodiments shown,the AC switching network 13A keeps the DC voltage under 60 V. In someembodiments, the AC switching network 13A is a special-purpose adiabaticswitched-capacitor network.

Once the AC switching network 13A has unfolded the AC voltage, aregulating circuit 16A, shown in FIG. 57, produces a final outputvoltage. In some embodiments, another switching network 16A between theAC switching network 13A and the regulating circuit 16A furtherconditions the voltage. If this is the case, then the caveats forseries-connected stages hold true since the AC switching network 13A isa special purpose switching network 12A. Some form of magnetic orelectric isolation is also common in AC-DC converters for safetyreasons. Hence, in FIG. 57, voltages: V_(AC), V_(DC), and V_(O) arepurposely defined as being agnostic to a common ground.

FIG. 58 shows an AC-DC converter corresponding to the architecture shownin FIG. 57. In this embodiment, the AC switching network 13A is asynchronous AC bridge rectifier followed by a reconfigurable two-phasestep-down cascade multiplier with three distinct conversion ratios (1/3,1/2, 1/1) while the regulating circuit 16A is a synchronous buckconverter. In operation, switches labeled 7 and 8 are always incomplementary states. During the positive portion of the AC cycle (0 toπ radians) all switches labeled “7” are closed while all switcheslabeled “8” are opened as shown in FIG. 59. Similarly, during thenegative portion of the AC cycle (π to 2π radians) all switches labeled8 are closed while all switches labeled “7” are opened as shown in FIG.60.

In addition to the inverting function provided by switches 7 and 8,switches 1A-1E and switches 2A-2E may be selectively opened and closedas shown in Table 1 to provide three distinct conversion ratios of: 1/3,1/2, and 1.

TABLE 1 V₂/V₁ 1A 1B 1C 1D 1E 2A 2B 2C 2D 2E 1/3 CLK CLK CLK CLK CLK CLKBCLKB CLKB CLKB CLKB 1/2 CLKB CLK CLK CLK CLK CLK CLKB CLKB CLKB CLKB 1/1ON ON ON OFF OFF ON ON ON OFF OFF

The AC switching network 13A is provided with a digital clock signalCLK. A second signal CLKB is also generated, which may simply be thecomplement of CLK (i.e. is high when CLK is low and low when CLK ishigh), or which may be generated as a non-overlapping complement. With aswitching pattern set in accordance with the first row of Table 1, theAC switching network 13A provides a step-down ratio of one-third (1/3).With a switching pattern set in accordance with the second row of Table1, the AC switching network 13A provides a step-down ratio of one-half(1/2). With a switching pattern set in accordance with the third row ofTable 1, the AC switching network 13A provides a step-down ratio of one.

Most power supplies attached to the wall meet some power factorspecification. Power factor is a dimensionless number between 0 and 1that defines a ratio of the real power flowing to apparent power. Acommon way to control the harmonic current and thus boost the powerfactor is by using an active power factor corrector. FIG. 61 shows anAC-DC converter 8 that controls harmonic current and boosts power factortowards unity. The illustrated AC-DC converter 8 features an ACswitching network 13A that receives an AC voltage from an AC source 14Band rectifies it. An output of the AC switching network 13A connects toan input of an active power-factor correction circuit 17A. The ACswitching network 13A may also provide voltage transformation via aswitched-capacitor circuit. The power-factor correction circuit 21Acontrols its input current so that it remains, to the greatest extentpossible, in-phase with the voltage waveform provided by the AC source14B. This drives reactive power toward zero. The output of thepower-factor correction circuit 17A is then provided to a regulatingcircuit 16A that operates in the same way as shown in FIG. 57.

FIG. 62 shows a particular embodiment of FIG. 55's modular DC-DCconverter 10 connected between first and second circuits 51, 52. Thefirst and second circuits 51, 52 can be a source, a load, or anothercircuit, such as a power converter, a PFC circuit, or an EMI filter.

The illustrated modular DC-DC converter 10 includes a regulating circuit16A, a switching network 12A, and an isolated controller 60. As usedherein, a circuit having an input and an output is considered isolatedif the input voltage and the output voltage do not share a commonground. Such isolation can be carried out by having the input voltagecorrespond to an input voltage of a transformer and having the outputvoltage corresponds to an output voltage of a transformer. In someembodiments, the regulating circuit 16A is isolated. In otherembodiments, it is the switching network 12A that is isolated. Althoughonly one of the foregoing is needed to consider the modular DC-DCconverter 10 as a whole isolated, there are also embodiments in whichboth the switching network 12A and the regulating circuit 16A areisolated.

In some embodiments, the switching network 12A is an unregulatedswitched-capacitor converter having a fixed voltage-conversion ratio.These embodiments generally include a regulating circuit 16A to regulatethe output of the switching network 12A. Examples of a suitableregulating circuit 16A include a boost converter, a buck converter, afly-back converter, and a linear regulator.

FIG. 63 shows a variation of the converter shown in FIG. 62 in which anLC filter 21A is added between the switching network 12A and the secondcircuit 52. The purpose of the LC filter is to promote adiabaticcharging of the switching network 12A via the method shown in FIG. 47.

FIG. 64 shows a particular embodiment of the modular DC-DC converter 10shown in FIG. 63. The regulating circuit 16A is implemented as afly-back converter having a switch S₁, a diode D₁, a capacitor C₁, and atransformer T₁. When operating in continuous conduction mode, theregulating circuit 16A transitions between first and second states. Inthe first state, the switch S₁ is closed, and the diode D₁ does notconduct. During this first state, the capacitor C₁ acts as a chargereservoir to supply power to the output of the regulator 16A. In thesecond state, the switch S₁ is opened and the diode D₁ conducts.

As shown in FIG. 64, the isolated controller 60 includes a first controlsignal CTR1 that controls the switching network 12A, a second controlsignal CTR2 that controls the regulating circuit 16A, and an isolationbarrier 61 between them. As a result, the first and second controlsignals CRT1, CTR2 have different grounds and connect to different sidesof the transformer T₁. The isolation barrier 61 can include any one ormore of sonic isolation, optical isolation, capacitive isolation,inductive isolation, and mechanical isolation.

The embodiment shown in FIG. 23 can be modified to operate with an ACsource 14B, as shown in FIG. 65, which shows a modular DC-DC converter10 connected between first and second circuits 51, 52. The modular DC-DCconverter 10 includes first and second switching networks 12A, 12B and aregulating circuit 16A. The first switching network 12A receives, at itsinput thereof, a voltage from the first circuit 51. The second switchingnetwork 12B provides its output to the second circuit 52. The regulatingcircuit 16A receives an output from the first switching network 12A andprovides its own output to an input of the second switching network 12B.An isolated controller 60 provides a first control signal to the firstswitching network 12A, a second control signal to the second switchingnetwork 12B, and a third control signal to the regulating circuit 16A.

Similarly, the embodiment shown in FIG. 26 can be modified to operatewith an AC source 14B, as shown in FIG. 66, which shows first and secondregulating circuits 16A, 16B and a switching network 12A. The firstregulating circuit 16A receives, at its input, a voltage from the firstcircuit 51. The second regulating circuit 16B provides its output to thesecond circuit 52. The switching network 12A receives an output from thefirst regulating circuit 16A and provides its own output to an input ofthe second regulating circuit 126. An isolated controller 60 provides afirst control signal to the first regulating circuit 16A, a secondcontrol signal to the regulating circuit 16B, and a third control signalto the switching network 12A. In some embodiments, as shown in FIG. 63,the second regulating circuit 16B can be implemented as an LC filter21A. The AC rectification stage 19A shown in FIG. 55 can be implementedin a variety of ways. In one embodiment, shown in FIG. 67, the rectifier19A features a fuse 71, a capacitor C₁, an AC bridge 80, and a firstelectromagnetic interference filter 70A between the AC bridge 80 and theAC source 14B. In another embodiment, shown in FIG. 68, a second EMIfilter 70B and a power-factor correction circuit 90 replaces thecapacitor C₁.

The first electromagnetic interference filter 70A, implementations ofwhich can be seen in FIGS. 69 and 70, reduces the common-mode anddifferential-mode noise produced by the AC-DC converter 8 by a desiredamount. The extent to which such noise is reduced is typically set by agovernment body, such as the FCC.

The AC bridge 80 accepts an AC voltage and outputs an average DCvoltage. A particular implementation of an AC bridge 80 is shown in FIG.71. The bridge includes first, second, third, and fourth diodes D₁, D₂,D₃, D₄. In operation, the AC bridge 80 transitions between first andsecond states. In the first state, the first and third diodes D₁, D₃ arereverse biased, and the second and fourth diodes are forward biased. Inthe second state, the second and fourth diodes D₂, D₄ are forward biasedand the first and third diodes D₁, D₃ are reverse biased.

Many modern devices require different voltages to operate differentcomponents, such as power management integrated circuits (PMICs) in cellphones. For example, one voltage may be required to operate a processor,whereas another voltage may be needed to operate a display. Inprinciple, one could have a separate transformation stage and regulationstage corresponding to each required output voltage. However, thissolution is wasteful both of physical space and of pin count. A solutionto this difficulty is that shown in FIG. 72, in which one transformationstage drives two or more regulation stages in parallel. Each regulationstage thus provides a separate output voltage. The regulator stage canbe any of those already described, including a linear regulator.

To ensure adiabatic charging of the switched-capacitor network in thetransformation stage, it is preferable that the majority of the powerdrawn by the various regulation stages come by way of a constantcurrent. This can be achieved, for example, by synchronizing theregulation stages so that they draw as constant a current as possible,thus avoiding larger resistive losses in the switched-capacitor networkof the transformation stage.

FIGS. 73-80 show specific implementations of modular power convertersthat conform to the architectural diagrams shown in FIGS. 22, 23, 24,and 26. In each implementation a regulating circuit or multipleregulating circuits may limit both the RMS charging current and the RMSdischarging current of at least one capacitor in each switching networkso all of these switching networks are adiabatically charged switchingnetworks. However, if decoupling capacitors 9A or 9B are present, thenthe ability of the regulating circuit to limit the RMS charging anddischarging current may be diminished. Capacitors 9A and 9B are optionaland to keep the output voltage fairly constant capacitor Co is used. Allof the stages share a common ground, however this need not be case. Forexample, if a regulating circuit is implemented as a fly-back converterthan the ground can be separated easily, even a switching network canhave separate grounds through capacitive isolation. Furthermore, forsimplicity, the switching network in each implementation has a singleconversion ratio. However, reconfigurable switching networks thatprovide power conversion at multiple distinct conversion ratios may beused instead.

In operation, switches labeled “1” and “2” are always in complementarystates. Thus, in a first switched-state, all switches labeled “1” areopen and all switches labeled “2” are closed. In a secondswitched-state, all switches labeled “1” are closed and all switcheslabeled “2” are opened. Similarly, switches labeled “3” are “4” are incomplementary states, switches labeled “5” are “6” are in complementarystates, and switches labeled “7” are “8” are in complementary states.Typically, the regulating circuits operate at higher switchingfrequencies than the switching networks. However, there is norequirement on the switching frequencies between and amongst theswitching networks and regulating circuits.

FIG. 73 shows a step-up converter corresponding to the architectureshown in FIG. 11. In this embodiment, the switching network 12A is atwo-phase step-up cascade multiplier with a conversion ratio of 1:3while the regulating circuit 16A is a two-phase boost converter. Inoperation, closing switches labeled 1 and opening switches 2 chargescapacitors C₃ and C₄ while discharging capacitors C₁ and C₂. Conversely,opening switches 1 and closing switches 2 charges capacitors C₁ and C₂while discharging capacitors C₃ and C₄.

FIG. 74 shows bidirectional step-down converter corresponding to thearchitecture shown in FIG. 22. In this embodiment, the switching network12A is a two-phase step-down cascade multiplier with a conversion ratioof 4:1 while the regulating circuit 16A is synchronous buck converter.In operation, closing switches 1 and opening switches 2 chargescapacitors C₁, C₂, and C₃ while discharging capacitors C₄, C₅, and C₆.Conversely, opening switches 1 and closing switches 2 charges capacitorsC₄, C₅, and C₆ while discharging capacitors C₁, C₂, and C₃. All of theactive components are implemented with switches so that the convertercan process power in both directions.

FIG. 75 shows a step-up converter consistent with the architecture shownin FIG. 24. In this embodiment, the regulating circuit 16A is boostconverter while the switching network 12A is a two-phase step-upseries-parallel switched-capacitor converter with a conversion ratio of1:2. In operation, closing switches 1 charges capacitor C₂ whiledischarging capacitor C₁. Closing switches 2 has the complementaryeffect.

FIG. 76 shows a bidirectional up-down converter consistent with thearchitecture shown in FIG. 24. In this embodiment, the regulatingcircuit 16A is synchronous four switch buck-boost converter while theswitching network 12A is a two-phase step-up cascade multiplier with aconversion ratio of 1:4. In operation, closing switches 1 chargescapacitors C₄, C₅, and C₆ while discharging capacitors C₁, C₂, and C₃.Closing switches 2 has the complementary effect. All of the activecomponents are implemented with switches so that the converter canprocess power in both directions.

FIG. 77 shows an inverting up-down converter consistent with thearchitecture shown in FIG. 2. In this embodiment, the first switchingnetwork 12A is a step-down series-parallel switched-capacitor converterwith a conversion ratio of 2:1, the first regulating circuit 16A is abuck/boost converter; and the second switching network 12B is a step-upseries-parallel switched-capacitor converter with a conversion ratio of1:2. In operation, closing switches 1 charges capacitor C₁ while closingswitches 2 discharges capacitor C₁. Similarly, closing switches 7discharges capacitor C₂ while closing switches 8 charges capacitor C₂.

FIG. 78 shows a bidirectional inverting up-down converter consistentwith the architecture shown in FIG. 23. In this embodiment, the firstswitching network 12A is a two-phase step-down series-parallelswitched-capacitor converter with a conversion ratio of 2:1, theregulating circuit 16A is a synchronous buck/boost converter and thesecond switching network 12B is a two-phase step-up series-parallelswitched-capacitor converter with a conversion ratio of 1:2. Inoperation, closing switches 1 charges capacitor C₁ while dischargingcapacitor C₂. Closing switches 2 has the complementary effect.Similarly, closing switches 7 charges capacitor C₄ while dischargingcapacitor C₃. Closing switches 2 has the complementary effect. All ofthe active components are implemented with switches so that theconverter can process power in both directions.

FIG. 79 shows a step-down converter consistent with the block diagramshown in FIG. 26. In this embodiment, the first regulating circuit 300Ais a boost converter, the switching network 200 is a two-phase step-upseries-parallel switched-capacitor converter with a conversion ratio of1:2, and the second regulating circuit 300B is a boost converter. Inoperation, closing switches 1 charges capacitors C₁ and C₂ whilesimultaneously discharging capacitors C₃ and C₄. Closing switches 2 hasthe complementary effect.

FIG. 80 shows a bidirectional up-down converter consistent with theblock diagram shown in FIG. 26. In this embodiment, the first regulatingcircuit 300A is a synchronous boost converter, the switching network 200is a two-phase fractional step-down series-parallel switched-capacitorconverter with a conversion ratio of 3:2 and the second regulatingcircuit 300B is a synchronous buck converter. In operation, closingswitches 1 charges capacitors C₃ and C₄ while simultaneously dischargingcapacitors C₁ and C₂. Closing switches 2 has the complementary effect.All of the active components are implemented with switches so that theconverter can process power in both directions.

It should be understood that the topology of the regulating circuit canbe any type of power converter with the ability to regulate the outputvoltage, including, but without limitation, synchronous buck,three-level synchronous buck, SEPIC, soft switched or resonantconverters. Similarly, the switching networks can be realized with avariety of switched-capacitor topologies, depending on desired voltagetransformation and permitted switch voltage.

The physical implementation of the foregoing switching networks 12Aincludes four primary components: passive device layers, active devicelayers, interconnect structures, and thru-vias. The passive devicelayers have passive devices, such as capacitors. The active devicelayers have active devices, such as switches.

The separation of active and passive devices in different layers arisesbecause active devices are made by CMOS processing. Thus, if one haspassive devices on the same layer, they must be made by CMOS-compatibleprocessing steps to avoid destroying the active devices. This constraintmakes it difficult to manufacture capacitors that provide highcapacitance in a small area of the chip. It also makes it difficult tomake high Q inductors. To avoid these difficulties, it is preferable toproduce integrated passive devices on their own wafer with a processflow that is optimized for producing such passive devices.

In some embodiments, the devices are integrated into a single monolithicsubstrate. In other embodiments, the devices are integrated intomultiple monolithic substrates. The monolithic substrates are typicallymade of semiconductor material, such as silicon.

In a preferred practice, one makes passive devices on a passive devicelayer using an integrated passive device process and makes activedevices on an active device layer using a CMOS process. These devicelayers are electrically connected together through a fine interconnectstructure that includes thru-vias to allow electrical connections acrossdevice layers.

FIG. 81 shows a circuit block diagram of a modular converter that usescapacitors in a switched-capacitor circuit to transfer energy. The blockdiagram shows a stack of layers that includes layers for both switchesand capacitors. The switches within the stack of layers include firstand second switches S₁, S₂. The capacitors within the stack of layersincludes first and second capacitors C₁, C₂. A discrete inductor L₁ ismounted outside the layer stack.

The layers within the stack of layers in FIG. 81 can be stacked indifferent ways. FIGS. 82-84 show side views of different ways ofstacking layers, and placement of the interconnect structure and viascorresponding to each such configuration of layers. The active devicelayers include switches while the passive device layers includecapacitors.

In FIG. 82, an active device layer connects to a printed-circuit boardvia a set of C4 bumps and a passive device layer is stacked above theactive device layer. Thru-vias TV provide a connection between theprinted-circuit board and an interconnect structure between the twolayers.

In FIG. 83, this orientation is reversed, with the passive layer beingconnected to the printed-circuit board by the C4 bumps and the activelayer above the passive layer. Once again, thru-vias TV provide aconnection between the printed-circuit board and an interconnectstructure between the two layers.

FIG. 84 shows the possibility of stacking multiple passive or activelayers. In the particular embodiment shown, there are n passive deviceslayers and one active device layer. Through vias TV provide a path forconnecting the printed-circuit board to interconnect structures betweenadjacent layers.

FIG. 85 shows an embodiment that has at least two device layers, one ofwhich has switches and another of which has capacitors.

The C4 bumps are laid out along the printed-circuit board at a firstpitch. An interconnect structure includes C5 bumps laid out at a secondpitch that is smaller than the first pitch. An example of such C5 bumpscan be seen in FIG. 95.

Each passive layer has capacitors that occupy a certain footprint on thechip. The capacitors are located such that each one is within afootprint of a switch on an active layer that is above or below thepassive layer. Such an arrangement helps reduce energy loss and otherparasitic losses in the interconnect structures.

Additional permutations arise because, as a result of the nature ofknown semiconductor fabrication processes, it is common to process onlyone face of a wafer. This face of the wafer has devices integrated intoit. For this reason, it is called the “device face.”

For each stack configuration, there are now additional permutationsconcerning whether the device face is an upper face or a lower face. Fora given layer, with reference to the z-axis shown in FIGS. 82-84, an“upper face” of that layer faces in the +z direction a “lower face”faces in the −z direction.

As used herein, a layer is said to “face” the +z direction if a vectorthat is perpendicular to a plane defined by that layer and that isdirected in a direction away from that layer is directed in the +zdirection. A layer is said to face in the −z direction if it does notface the +z direction.

For the case in which there are only two device layers, FIGS. 86-88 showthe four possible configurations of device faces when the upper layer isthe passive layer, as shown in FIG. 82. FIGS. 90-93 show the fourpossible configurations of device faces when the upper layer is theactive layer, as shown in FIG. 83.

In FIG. 86, the active layer's device face is its upper face and thepassive layer's device face is its lower face. Given that there are onlytwo layers, this means they face each other. FIG. 88 shows a conversecase in which the passive layer's device face is its upper face and theactive layer's device face is its lower face. In FIG. 87, both thedevice faces of both the active and passive layers are on upper faces,whereas in FIG. 89 both are on lower faces.

FIGS. 90-93 show the converse of FIGS. 86-89 for the case in which theactive layer is now the upper layer. In FIG. 90, the active devices areon a lower face and the passive devices are on an upper face. Sincethere are only two layers, the active and passive devices face eachother as they did in FIG. 86. In FIG. 91, the active devices and passivedevices are on upper faces of their respective layers, whereas in FIG.93 they are on lower faces of their respective layers. In FIG. 92, theactive devices are on an upper face and the passive devices are on alower face.

Naturally, certain configurations are preferable to others. The choicewill depend upon numerous factors, most of which relate to thru-viatechnology and the number of pins that are available to connect thelayers to external circuitry.

The passive device layer and active device layer can be in any form whenattached. Two common choices would be in die or wafer form.

FIGS. 94-95 show cross-sections of two die-to-die arrangements in whichan interconnect structure connects switches in an active die tocapacitors on a passive die. In FIG. 94, the switches connect to aplanar capacitor whereas in FIG. 95 the switches connect to a trenchcapacitor. The first bumps C4, which provide the electrical connectionsfrom the die stack to the printed-circuit board, and through-vias TV areomitted in FIGS. 94-95 but can be seen in FIGS. 96-97.

Although any kind of capacitor can be used, trench capacitors arepreferable to planar capacitors because trench capacitors offer greatercapacitance per unit of die area than planar capacitors, sometimes byone or two orders of magnitude. Additionally, trench capacitors offerlower equivalent series resistance than planar capacitors. Both of thesecapacitor attributes are desirable for use in power converters that usecapacitive energy transfer because they affect the efficiency of thepower converter.

As shown in FIGS. 94-95, an interconnect structure connects the switcheson the active die to the capacitors on the passive die. Thisinterconnect structure can be implemented in numerous ways. In the caseof FIGS. 94-95, the interconnect structure is the union of a multilayerinterconnect structure on the passive die, a single layer of secondbumps C5, and a multilayer interconnect structure on the active die. Theonly requirements are that the interconnect structure connects theswitches on one device layer to the capacitors on the other devicelayer, that the two device layers are stacked one on top of the other,and that the second bumps C5 have a much finer pitch than the firstbumps C4. In some embodiments, the pitch of the second bumps C5 is fourtimes greater than the pitch of the first bumps. As used herein, “pitch”means the number of bumps per unit length.

FIGS. 96-97 show another embodiment implemented by wafer-to-waferstacking. In this embodiment, there is no need for the second bumps C5.Instead, the active and passive wafers electrically connect to eachother using a bonding process. In FIG. 96, the device face of the activelayer is its lower face and in FIG. 97, the device face of the activelayer is its upper face. Examples of suitable bonding processes arecopper-copper and oxide-oxide bonding. Furthermore, FIGS. 96-97 show thethru-vias and some of the first bumps C4, which were omitted in FIGS.94-95.

A switched-capacitor power converter of the type discussed herein has agreat many switches and capacitors in a switched-capacitor powerconverter. These all have to be interconnected correctly for the powerconverter to operate. There are many ways to physically lay out theconducting paths that interconnect these components. However, not all ofthese ways are equally efficient. Depending on their geometry, some ofthese conducting paths may introduce noticeable parasitic resistanceand/or inductance. Because there are so many interconnections, it can bea daunting challenge to choose a set of interconnections that will bothprovide acceptable parasitic resistance and inductance for the powerconverter as a whole.

One method that can be used to control these parasitic quantities is topartition the switches and capacitors.

One way to reduce such parasitic quantities is to choose the shape andlocations of the switches on the active layer so that they fit beneaththe capacitors on the passive layer. This avoids forcing current toundertake a long journey along the faces of the layers as it travelsbetween a switch and a capacitor. An example of this technique is shownin FIG. 99, in which eight switches S₁-S₈ and a controller 20A aredisposed on an active layer that is located below a passive layer havingtwo capacitors. Although the switches are not completely visible throughthe passive layer, their locations are marked by dotted lines on FIG.99. The figure shows a first capacitor C₁ on top of switches S₁, S₂, S₅,S₆ and a second capacitor C₂ on top of switches S₃, S₄, S₇, S₈.

Another way to reduce such parasitic quantities arises from recognizingthat switches in a switching network 12A are usually active devices thatare implemented with transistors. The switching network 12A may beintegrated on a single monolithic semiconductor substrate or on multiplemonolithic semiconductor substrates, or formed using discrete devices.Furthermore, since the device is a power converter, each switch may beexpected to carry a large amount of current. A switch that carries agreat deal of current is often implemented by numerous current pathsconnected in parallel to a common terminal.

In a switch as described above, the current paths that make up theswitch are physically located side-by-side and thus occupy a spacehaving a non-zero width. These current paths all connect to a terminalthat is itself connected to a conducting path. An example of thisconfiguration is shown in FIG. 98 and FIG. 101. In particular, FIG. 101shows a transistor on a first layer and a capacitor on a lower layer.The transistor has first, second, and third current paths with thesecond current path being between the first and third. The three currentpaths extend between one source terminal and one drain terminal of thetransistor.

Some current entering the source terminal shown in FIG. 101 goesstraight ahead into the second current path. But some of it turns leftor right before turning again to proceed down the first and thirdcurrent paths. At the other end of the transistor's channel, currentthat traversed the first and third current paths must again make a turnto reach the drain terminal. These currents are referred to as “lateral”current.

Similarly, the lower layer of FIG. 101 shows a capacitor that has threeseparate current paths connected to first and second capacitorterminals. In the course of being charged and discharged, some lateralcurrent is inevitable for reasons discussed in connection with thetransistor in the upper layer.

One way to reduce this lateral current is to partition the switches andthe capacitors into numerous partitions, as shown in FIG. 98 and FIG.102. This partitioning essentially involves converting an n-terminaldevice into an (n+m) terminal device where m depends on the number ofpartitions. Thus, after having been partitioned, the two-terminalcapacitor of FIG. 101 is transformed into a six-terminal capacitor inFIG. 102. Similarly, the source terminal and drain terminal of thetransistor in FIG. 101 is transformed into three source terminals andthree drain terminals in the transistor of FIG. 102.

The difference between FIGS. 101 and 102 is that each current path inFIG. 102 has its own terminal. In contrast, in FIG. 101, all currentpaths share the same terminals. Thus, FIG. 101 shows three current pathsconnected in parallel, whereas FIG. 102 shows three current paths thatare partitioned and therefore isolated from each other.

The three current paths shown collectively represent a switch on anactive layer that is formed by various doping profiles along a piece ofsilicon to provide charge carriers and then connecting those three linesto a pair of external terminals, as shown in FIG. 101, or connectingeach line to its own pair of external terminals, as shown in FIG. 102.

The capacitor represented by the lower layer of FIG. 101 is atwo-terminal capacitor like any conventional capacitor. Prior artconverters use capacitors of this type. However, unlike prior artconverters, which use two-terminal capacitors, a converter as disclosedherein uses a six-terminal capacitor as shown FIG. 102. Although such acapacitor is more complex because it has more terminals that need to beboth made and properly aligned, it reduces parasitic effects caused bylateral current.

Similarly, the transistor switch represented by the upper layer of FIG.101 has one source terminal and one drain terminal. This is the kind oftransistor that is used in conventional power converters. In contrast,the transistor represented by the upper layer of FIG. 102 has threesource terminals and three drain terminals. Although such a transistoris more complex because it has more terminals that need to be both madeand properly aligned, it reduces parasitic effects caused by lateralcurrent.

It should be apparent that the act of partitioning isgeometry-independent. Its essence is that of turning an n-terminaldevice into an (n+m) terminal device in an effort to reduce parasiticeffects. There is no requirement that the device be oriented in anyparticular way. In particular, there is no requirement that thepartitioning be carried out in only one dimension as shown in FIG. 102.For example, it is quite possible to partition a component along x and ydirections as shown in the nine-partition switch of FIG. 100 and thesix-partition capacitor shown in FIG. 103.

Both the techniques shown in FIG. 102 and FIG. 103 reduce the verticaland lateral distance between the active and passive devices while alsoproviding a uniform current distribution to each individual switchand/or switched-capacitor cell. This tends to reduce the parasiticresistance and inductance of the connection between the switches andcapacitors. This offers considerable advantages. Parasitic inductancelimits the switching speed while parasitic resistance limits theefficiency of the power conversion process.

Among other advantages, the arrangements described above avoid thecomponent and pin count penalty, reduce the energy loss in the parasiticinterconnect structures, and reduces the total footprint of powerconverters that use capacitors to transfer energy.

Switching networks along the lines of the foregoing can be used tocontrol a power converter in a travel adapter 13, as shown in FIG. 104.Such a travel adapter 13 outputs a DC voltage at a USB port 15 thereof.

In some implementations, a computer accessible storage medium includes adatabase representative of one or more components of the converter. Forexample, the database may include data representative of a switchingnetwork that has been optimized to promote low-loss operation of acharge pump.

Generally speaking, a computer accessible storage medium may include anynon-transitory storage media accessible by a computer during use toprovide instructions and/or data to the computer. For example, acomputer accessible storage medium may include storage media such asmagnetic or optical disks and semiconductor memories.

Generally, a database representative of the system may be a database orother data structure that can be read by a program and used, directly orindirectly, to fabricate the hardware comprising the system. Forexample, the database may be a behavioral-level description orregister-transfer level (RTL) description of the hardware functionalityin a high level design language (HDL) such as Verilog or VHDL. Thedescription may be read by a synthesis tool that may synthesize thedescription to produce a netlist comprising a list of gates from asynthesis library. The netlist comprises a set of gates that alsorepresent the functionality of the hardware comprising the system. Thenetlist may then be placed and routed to produce a data set describinggeometric shapes to be applied to masks. The masks may then be used invarious semiconductor fabrication steps to produce a semiconductorcircuit or circuits corresponding to the system. In other examples,Alternatively, the database may itself be the netlist (with or withoutthe synthesis library) or the data set.

Having described one or more preferred embodiments, it will be apparentto those of ordinary skill in the art that other embodimentsincorporating these circuits, techniques and concepts may be used.Accordingly, it is submitted that the scope of the patent should not belimited to the described embodiments, but rather, should be limited onlyby the spirit and scope of the appended claims.

The invention claimed is:
 1. An apparatus to convert a first voltageinto a second voltage, the apparatus comprising: an inverter to providean AC signal to a bridge rectifier to be coupled to the inverter viamagnetically coupled coils to facilitate an inter-capacitor chargetransfer within a switched-capacitor arrangement based, at least inpart, on one or more switching patterns of one or more switches of aplurality of switches to be controllable to switch between the switchingpatterns according to one or more switching frequencies, the inverter tobe coupled to a primary coil of the magnetically coupled coils and thebridge rectifier to be coupled to a secondary coil of the magneticallycoupled coils, wherein the switched-capacitor arrangement to output atleast a DC signal based, at least in part, on the inter-capacitor chargetransfer.
 2. The apparatus of claim 1, wherein the DC signal to includea DC voltage to be regulated via at least one voltage regulator so as toproduce a regulated voltage, the regulated voltage to be substantiallythe same as the second voltage.
 3. The apparatus of claim 2, wherein theregulated voltage to be produced based, at least in part, on at leastone inductor operable to at least partially constrain a rate of theinter-capacitor charge transfer within the switched-capacitorarrangement.
 4. The apparatus of claim 1, wherein the bridge rectifiercomprises a half-bridge rectifier; a full bridge rectifier; anasynchronous bridge rectifier; and/or a synchronous bridge rectifier. 5.The apparatus of claim 1, wherein the plurality of switches to becontrollable via a controller.
 6. The apparatus of claim 5, wherein thecontroller comprises an isolated controller.
 7. The apparatus of claim1, wherein the inverter to be implemented, at least in part, in a cellphone.
 8. An apparatus to convert a first voltage into a second voltage,the apparatus comprising: a charge pump including two or more capacitorsinterconnected with one or more switches of a plurality of switches soas to implement a charge transfer between the two or more capacitors,the charge transfer to be implemented based, at least in part, on arectified AC voltage to be provided via a rectifier to be magneticallycoupled to an inverter; wherein the rectifier to be magnetically coupledto the inverter via a pair of electrical coils, the inverter to becoupled to a first coil of the electrical coils and the rectifier to becoupled to a second coil of the electrical coils, and wherein the chargepump to generate an output voltage for a regulator based, at least inpart, on the charge transfer between the two or more capacitors.
 9. Theapparatus of claim 8, wherein the second coil comprises a primary coiland the first coil comprises a secondary coil.
 10. The apparatus ofclaim 8, wherein the regulator to generate a regulated voltage to besubstantially the same as the second voltage.
 11. The apparatus of claim10, wherein the regulated voltage to be generated based, at least inpart, on an inductance to constrain the charge transfer between the twoor more capacitors via at least one of the one or more switches of theplurality of switches.
 12. The apparatus of claim 8, wherein theplurality of switches to be controllable via a controller.
 13. Theapparatus of claim 12, wherein the controller comprises an isolatedcontroller.
 14. The apparatus of claim 12, wherein the controller tohave separate grounds.
 15. The apparatus of claim 8, wherein the chargepump to be implemented, at least in part, in a cell phone.
 16. Anapparatus to convert a first voltage into a second voltage, theapparatus comprising: a regulator to generate a regulated voltage based,at least in part, on an output voltage to be outputted by a switchedcapacitor arrangement based, at least in part, on an input voltage, theregulator comprising an indictor to be arranged in an electricalconfiguration with the switched capacitor arrangement so as constrain arate of an inter-capacitor charge transfer between two or morecapacitors of the switched capacitor arrangement based, at least inpart, on an operation of one or more switches of a plurality ofswitches, wherein the input voltage to be provided via an AC voltagerectifier to be arranged in a magnetic configuration with an inverter,the magnetic configuration to include at least a pair of magneticallycoupled coils, wherein the inverter to be coupled to a primary coil ofthe magnetically coupled coils and the AC voltage rectifier to becoupled to a secondary coil of the magnetically coupled coils.
 17. Theapparatus of claim 16, wherein the output voltage to comprise anintermediate voltage.
 18. The apparatus of claim 16, wherein theregulated voltage to be substantially the same as the second voltage.19. The apparatus of claim 16, wherein the regulator comprises a buckconverter; a boost converter; a fly-back converter; a buck-boostconverter; a forward converter; a half-bridge converter; a full-bridgeconverter; a Cuk converter; a resonant converter; and/or a linearregulator.
 20. The apparatus of claim 16, wherein the regulator to beimplemented, at least in part, in a cell phone.